Response adjustable temperature sensor for transponder

ABSTRACT

A response-adjustable temperature sensor ( 306 , Rext), particularly for a transponder ( 102, 200, 400 ) capable of measuring one or more parameters (e.g., temperature, pressure) and transmitting a data stream (FIGS.  3 C,  4 B) to an external reader/interrogator ( 106 ). The transponder typically operates in a passive mode, deriving its power (Vxx, Vcc, Vdd) from an RF interrogation signal received by an antenna system ( 210, 410 ), but can also operate in a battery-powered active mode. The transponder includes memory ( 238, 438 ) for storing measurements, calibration data, programmable trim settings ( 436   b ), transponder ID and the like. The temperature sensor is a temperature measurement device characterized by a resistance (Rext,  216, 416, 716 ) and a temperature sensing circuit ( 306 ) comprising a temperature sensing transistor (Q 1 ) which exhibits a predictable change in its base-emitter voltage due to temperature, and transistors (P 1 , N 1 , P 2 , N 2 ) connected for mirroring a current (I(T)) through the temperature sensing transistor and through the resistance. The resistance may be a fixed, temperature-independent resistor or a resistance that has a resistance value that predictably varies with temperature, such as a thermistor ( 716 ). Temperature response and resolution can be adjusted while maintaining desired current levels by varying the value of the resistance and also by selecting a thermistor with a suitable temperature coefficient. User-settable trimming bits can be used to determine scaling of the transponder output. Replacing an external pressure-sensing capacitor with a fixed-value capacitor allows further flexibility in scaling of the output when only temperature readings are needed.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application relates to copending PCT application Ser. No.PCT/US99/29723 entitled POWER ON RESET FOR TRANSPONDER; copending PCTapplication Ser. No. PCT/US99/29827 entitled PROGRAMMABLE MODULATIONINDEX FOR TRANSPONDER; and copending PCT application Ser. No.PCT/US99/29840 entitled PROGRAMMABLE TRIMMING FOR TRANSPONDER, all filedDec. 15, 1999.

This application is a continuation-in-part of commonly-owned, copendingPCT application Ser. No. PCT/US99/29890 entitled RELAXATION OSCILLATORFOR TRANSPONDER, filed Dec. 15, 1999, which in turn claims the benefitof U.S. Provisional Patent Application No. 60/134,455, filed May 17,1999 by Yones.

TECHNICAL FIELD OF THE INVENTION

The present invention relates to temperature sensors and, moreparticularly, temperature sensors in conjunction with transponders formeasuring and transmitting pressure and temperature measurements to anexternal receiver (reader, or reader/interrogator) and, moreparticularly, for temperature response adjustment for the temperaturesensors.

BACKGROUND OF THE INVENTION

Safe, efficient and economical operation of a motor vehicle depends, toa significant degree, on maintaining correct air pressure in all (each)of the tires of the motor vehicle. Operating the vehicle with low tirepressure may result in excessive tire wear, steering difficulties, poorroad-handling, and poor gasoline mileage, all of which are exacerbatedwhen the tire pressure goes to zero in the case of a “flat” tire.

The need to monitor tire pressure when the tire is in use is highlightedin the context of “run-flat” (driven deflated) tires, tires which arecapable of being used in a completely deflated condition. Such run-flattires, as disclosed for example in commonly-owned U.S. Pat. No.5,368,082, incorporated in its entirety by reference herein, mayincorporate reinforced sidewalls, mechanisms for securing the tire beadto the rim, and a non-pneumatic tire (donut) within the pneumatic tireto enable a driver to maintain control over the vehicle after acatastrophic pressure loss, and are evolving to the point where it isbecoming less and less noticeable to the driver that the tire has becomedeflated. The broad purpose behind using run-flat tires is to enable adriver of a vehicle to continue driving on a deflated pneumatic tire fora limited distance (e.g., 50 miles, or 80 kilometers) prior to gettingthe tire repaired, rather than stopping on the side of the road torepair the deflated tire. Hence, it is generally desirable to provide alow tire pressure warning system within in the vehicle to alert (e.g.,via a light or a buzzer) the driver to the loss of air pressure in apneumatic tire.

To this end, a number of electronic devices and systems are known formonitoring the pressure of pneumatic tires, and providing the operatorof the vehicle with either an indication of the current tire pressure oralerting the operator when the pressure has dropped below apredetermined threshold level.

For example, U.S. Pat. No. 4,578,992 (Galasko, et al; 04/86),incorporated in its entirety herein, discloses a tire pressureindicating device including a coil and a pressure-sensitive capacitorforming a passive oscillatory circuit having a natural resonantfrequency which varies with tire pressure due to changes caused to thecapacitance value of the capacitor. The circuit is energized by pulsessupplied by a coil positioned outside the tire and secured to thevehicle, and the natural frequency of the passive oscillatory circuit isdetected. The natural frequency of the coil/capacitor circuit isindicative of the pressure on the pressure-sensitive capacitor.

It is also known to monitor tire pressure with an electronic device thatis not merely a passive resonant circuit, but rather is capable oftransmitting a radio frequency (RF) signal indicative of the tirepressure to a remotely-located receiver. Such a “transmitting device”may have its own power supply and may be activated only when thepressure drops below a predetermined threshold. Alternatively, thetransmitting device may be activated (“turned ON”) by an RF signal fromthe remotely-located receiver, in that case the receiver is consideredto be an “interrogator”. Additionally, the transmitting device may bepowered by an RF signal from the interrogator. Additionally, theelectronic device which monitors the tire pressure may have thecapability of receiving information from the interrogator, in which casethe electronic device is referred to as a “transponder”.

As used herein, a “transponder” is an electronic device capable ofreceiving and transmitting radio frequency signals, and impressingvariable information (data) in a suitable format upon the transmittedsignal indicative of a measured condition (e.g., tire pressure) orconditions (e.g., tire pressure, temperature, revolutions), as well asoptionally impressing fixed information (e.g., tire ID) on thetransmitted signal, as well as optionally responding to informationwhich may be present on the received signal. The typical condition ofparamount interest for pneumatic tires is tire pressure. “Passive”transponders are transponders powered by the energy of a signal receivedfrom the interrogator. “Active” transponders are transponders havingtheir own power supply (e.g., a battery), and include activetransponders that remain in a “sleep” mode, using minimal power, until“woken up” by a signal from an interrogator, or by an internal periodictimer, or by an attached device. As used herein, the term “tag” referseither to a transponder having transmitting and receiving capability, orto a device that has only transmitting capability. Generally, tags thatare transponders are preferred in the system of the present invention.As used herein, the term “tire-pressure monitoring system” (TPMS)indicates an overall system comprising tags within the tires and areceiver that may be an interrogator disposed within the vehicle.

It is known to mount a tag, and associated condition sensor (e.g.,pressure sensor) within each tire of a vehicle, and to collectinformation from each of these transponders with a common singleinterrogator (or receiver), and to alert a driver of the vehicle to alow tire pressure condition requiring correction (e.g., replacing thetire). For example, U.S. Pat. No. 5,540,092 (Handfield, et al.; 1996),incorporated in its entirety by reference herein, discloses a system andmethod for monitoring a pneumatic tire. FIG. 1 therein illustrates apneumatic tire monitoring system (20) comprising a transponder (22) anda receiving unit (24).

Examples of RF transponders suitable for installation in a pneumatictire are disclosed in U.S. Pat. No. 5,451,959 (Schuermann; 09/95), U.S.Pat. No. 5,661,651 (Geschke, et al.; 08/97), and U.S. Pat. No. 5,581,023(Handfield, et al.; 12/96), all incorporated in their entirety byreference herein. The described transponder systems includeinterrogation units, pressure sensors and/or temperature sensorsassociated with the transponder, and various techniques for establishingthe identity of the tire/transponder in multiple transponder systems. Inmost cases, such transponders require battery power.

In some instances, a transponder may be implemented as an integratedcircuit (IC) chip. Typically, the IC chip and other components aremounted and/or connected to a substrate such as a printed circuit board(PCB).

Some proposed systems have relatively complex transponder-sensorcapabilities, including measurement and reporting of tire rotations andspeed, along with tire ID, temperature, and pressure. For example: U.S.Pat. No. 5,562,787 (Koch, et al.; 1996), and U.S. Pat. No. 5,731,754(Lee, Jr., et al.; 1998), incorporated in their entirety by referenceherein.

TRANSPONDER ENVIRONMENTAL CONSIDERATIONS

The environment within which a tire-mounted transponder must reliablyoperate, including during manufacture and in use, presents numerouschallenges to the successful operation of the transducer. For example,the sensors (e.g., pressure, temperature) used with the transponderpreferably will have an operating temperature range of up to 125° C.,and should be able to withstand a manufacturing temperature ofapproximately 177° C. For truck tire applications, the pressure sensormust have an operating pressure range of from about 50 psi to about 120psi (from about 345 kPa to about 827 kPa), and should be able towithstand pressure during manufacture of the tire of up to about 400 psi(about 2759 kPa). The accuracy, including the sum of all contributors toits inaccuracy, should be on the order of plus or minus 3% of fullscale. Repeatability and stability of the pressure signal should fallwithin a specified accuracy range.

However it is implemented, a tire transponder (tag) must therefore beable to operate reliably despite a wide range of pressures andtemperatures. Additionally, a tire transponder must be able to withstandsignificant mechanical shocks such as may be encountered when a vehicledrives over a speed bump or a pothole.

A device which can be used to indicate if a transponder or the tire hasbeen exposed to excessive, potentially damaging temperatures is the“MTMS” device or Maximum Temperature Memory Switch developed by. Prof.Mehran Mehregany of Case Western Reserve University. It is amicro-machined silicon device that switches to a closed state at acertain high-temperature point. The sensor switches from an “open” highresistance state of, for example, over 1 mega-ohm to a “closed” lowresistance state of, for example, less than 100 ohm.

Although it is generally well known to use pressure transducers inpneumatic tires, in association with electronic circuitry fortransmitting pressure data, these pressure-data systems for tires havebeen plagued by difficulties inherent in the tire environment. Suchdifficulties include effectively and reliably coupling RF signals intoand out of the tire, the rugged use the tire and electronic componentsare subjected to, as well as the possibility of deleterious effects onthe tire from incorporation of the pressure transducer and electronicsin a tire/wheel system. In the context of “passive” RF transponders thatare powered by an external reader/interrogator, another problem isgenerating predictable and stable voltage levels within the transponderso that the circuitry within the transponder can perform to its designspecification.

Suitable pressure transducers for use with a tire-mounted transponderinclude:

(a) piezoelectric transducers;

(b) piezoresistive devices, such as are disclosed in U.S. Pat. No.3,893,228 (George, et al.; 1975) and in U.S. Pat. No. 4,317,216 (Gragg,Jr.; 1982);

(c) silicon capacitive pressure transducers, such as are disclosed inU.S. Pat. No. 4,701,826 (Mikkor; 1987), U.S. Pat. No. 5,528,452 (Ko;1996), U.S. Pat. No. 5,706,565 (Sparks, et al.; 1998), andPCT/US99/16140 (Ko, et al.; filed Jul. 7, 1999);

(d) devices formed of a variable-conductive laminate of conductance ink;and

(e) devices formed of a variable-conductance elastomeric composition.

THE EFFECT OF TEMPERATURE ON GAS PRESSURE

In a broad sense, for a mass of any gas in a state of thermalequilibrium, pressure P, temperature T, and volume V can readily bemeasured. For low enough values of the density, experiment shows that(1) for a given mass of gas held at a constant temperature, the pressureis inversely proportional to the volume (Boyle's law), and (2) for agiven mass of gas held at a constant pressure, the volume is directlyproportional to the temperature (law of Charles and Gay-Lussac). Thisleads to the “equation of state” of an ideal gas, or the “ideal gaslaw”:

PV=μRT

where:

μis the mass of the gas in moles; and

R is a constant associated with the gas.

Thus, for a contained (fixed) volume of gas, such as air containedwithin a pneumatic tire, an increase in temperature (T) will manifestitself as an increase in pressure (P).

Because of the ideal gas law relationship, it is recognized that in thecontext of pneumatic tires, one problem that arises during operation oftire pressure sensors of any kind is that tires heat up as they are runfor longer periods of time. When a tire heats up, air that is confinedwithin the essentially constant and closed volume of the tire expands,thus causing increased pressure within the tire, though the overallamount of air within the tire remains the same. Since the pressurenominally is different, a tire pressure sensor can provide differentpressure readings when a tire is hot than would be the case if the tirewere cold. This is why tire and vehicle manufacturers recommend thatowners check their tire pressure when the tire is cold. Of course, witha remote tire pressure sensor, an operator may receive a continuousindication of tire pressure within the vehicle, but the indication maybe inaccurate because of the temperature change. Thus, it is necessaryto compensate for changes in temperature of the inflating medium (“gas”or air) within the pneumatic tire.

Patents dealing in one way or another with gas law effects in pneumatictires include: U.S. Pat. No. 3,596,509 (Raffelli; 1971), U.S. Pat. No.4,335,283 (Migrin; 1982), U.S. Pat. No. 4,126,772 (Pappas, et al.;1978), U.S. Pat. No. 4,909,074 (Gerresheim, et al.; 1990), U.S. Pat. No.5,050,110 (Rott; 1991), U.S. Pat. No. 5,230,243 (Reinecke; 1993), U.S.Pat. No. 4,966,034 (Bock, et al.; 1990), U.S. Pat. No. 5,140,851(Hettrich, et al.; 1992), U.S. Pat. No. 4,567,459 (Folger, et al.;1986), all of which are incorporated in their entirety by referenceherein.

U.S. Pat. No. 4,893,110 (Hebert; 1990), incorporated in its entirety byreference herein, discloses a tire monitoring device using pressure andtemperature measurements to detect anomalies. As mentioned therein, aratio of temperature and pressure provides a first approximation of anumber of moles of gas in the tire, which should remain constant barringa leak of inflation fluid from the tire. (column 1, lines 18-26). Moreparticularly, on each wheel are installed sensors (4) for pressure andsensors (6) for temperature of the tire, as well as elements (8 and 10)for transmitting the measured values as coded signals to a computer (12)on board the vehicle, such as disclosed in the aforementioned U.S. Pat.No. 4,703,650. The computer processes the measured values for pressureand temperature for each tire, and estimates for thepressure/temperature ratio (P/T estimate) are calculated for each wheel.Generally, the ratio for one of the tires is compared with the ratio forat least another one of the tires, and an alarm is output when a result(N) of the comparison deviates from a predetermined range of values.

TECHNIQUES FOR TRANSMITTING PRESSURE AND TEMPERATURE READINGS FROM ATIRE

Given that pressure and temperature conditions within a pneumatic tirecan both be measured, various techniques have been proposed to transmitsignals indicative of the measured pressure and temperature conditionsto an external interrogator/receiver. For example, the following patentsare incorporated in their entirety by reference herein:

transmit the signals individually, distinguished by phase displacements:U.S. Pat. No. 4,174,515 (Marzolf; 1979);

multiplex the signals: U.S. Pat. No. 5,285,189 (Nowicki, et al.; 1994),U.S. Pat. No. 5,297,424 (Sackett; 1994);

encoding the signals as separate segments of a data word: U.S. Pat. No.5,231,872 (Bowler, et al.; 1993), and U.S. Pat. No. 4,695,823 (Vernon;1987) which also incorporates both the telemetry and the pressure and/ortemperature sensors on the same integrated circuit chip;

transmission between coils mounted on the wheel and on the vehicle: U.S.Pat. No. 4,567,459 (Folger, et al.; 1986);

use a frequency-shift key (FSK) signal: U.S. Pat. No. 5,228,337 (Sharpe,et al.; 1993);

backscatter-modulate the RF signal from the interrogator with the tirecondition parameter data from the sensors, then return the backscattermodulated signal to the interrogator: U.S. Pat. No. 5,731,754 (Lee, Jr.,et al.; 1998).

U.S. Pat. No. 4,703,650 (Dosjoub, et al.; 1987), incorporated in itsentirety by reference herein, discloses a circuit for coding the valueof two variables measured in a tire, and a device for monitoring tiresemploying such a circuit. The coding circuit comprises an astablemultivibrator which transforms the measurement of the variables, forinstance pressure and temperature, into a time measurement. The astablemultivibrator delivers a pulse signal whose pulse width is a function ofthe temperature and the cyclic ratio of which is a function of thepressure.

U.S. Pat. No. 5,054,315 (Dosjoub; 1991), incorporated in its entirety byreference herein, discloses a technique for coding the value of severalquantities measured in a tire. As disclosed therein:

“Coding of the value of any number of quantities measured in a tire, forexample its pressure and its temperature, is carried out using a ratioof time intervals TP/Tr, Tt/Tr. This frees the device from the effect ofthe time shift of the modulation system, the time shift affectingsimultaneously the numerator and the denominator of said ratio.”(Abstract)

SUMMARY OF THE INVENTION

According to the invention, a temperature measurement device isdisclosed which comprises a resistance, and a temperature sensingcircuit comprising a temperature sensing transistor which exhibits apredictable change in its base-emitter voltage due to temperature, andtransistors connected for mirroring a current through the temperaturesensing transistor and through the resistance.

According to the invention, the resistance is a fixed resistor or aresistance that has a resistance value that predictably varies withtemperature, such as a thermistor, for example. The fixed resistor has aresistance value that is substantially independent of temperature, andwhich resistance value may be between about 20.5 kilohms and about 455kilohms, or which alternatively may be greater than about 200 kilohms.

According to the invention, within a desired temperature measurementrange, the resistance has a resistance value great enough to preventunacceptable levels of the current.

According to the invention, there is disclosed a method of adjustingtemperature response for a temperature sensing circuit connected to avoltage-to-current converting resistance wherein the temperature sensingcircuit includes a temperature sensing transistor which exhibits apredictable change in its base-emitter voltage due to temperature, andtransistors connected for mirroring a current through the temperaturesensing transistor and through the resistance. This method ischaracterized by the step of selecting the resistance value in order toproduce a desired slope for the temperature response.

According to the invention, the method includes utilizing a fixedresistor for the resistance, or alternatively utilizing a resistancethat predictably varies with temperature.

According to an embodiment of the invention utilizing a resistance whichpredictably varies with temperature, the method includes selecting atemperature coefficient for the resistance which has a value greatenough compared to the temperature coefficient of the temperaturesensing transistor in order to increase the slope of the temperatureresponse compared to the slope which would result from utilizing aminimum resistance valued fixed resistor for the resistance; andselecting a nominal value for the resistance such that the resistancevalues are great enough to prevent unacceptable levels of the currentwithin a desired temperature measurement range.

According to another embodiment of the invention utilizing a resistancewhich predictably varies with temperature, the method includes selectinga temperature coefficient for the resistance which counterbalances thetemperature coefficient of the temperature sensing transistor in orderto produce an approximately zero slope of the temperature responsewithin a portion of a desired temperature measurement range; andselecting a nominal value for the resistance such that thetemperature-varying resistance value is great enough to preventunacceptable levels of the current within the desired temperaturemeasurement range.

According to the invention, an RF transponder is disclosed whichcomprises a resistance; a temperature sensing circuit comprising atemperature sensing transistor which exhibits a predictable change inits base-emitter voltage due to temperature, and transistors connectedfor mirroring a temperature-indicative current through the temperaturesensing transistor and through the resistance; circuitry for convertingthe mirrored current to a temperature reading which is proportional tothe mirrored current; and a value for the resistance which predictablyvaries with temperature, such as, for example, a thermistor.

According to the invention, the RF transponder is further characterizedin that: within a desired temperature measurement range, the resistancehas resistance values great enough to prevent unacceptable levels of themirrored current. The RF transponder may be further characterized inthat the resistance has a temperature coefficient great enough, relativeto the temperature coefficient of the temperature sensing transistor, toincrease the slope of the temperature reading versus temperaturecompared to the slope which would result from utilizing a minimumresistance valued fixed resistor for the resistance. Alternatively, theRF transponder is further characterized in that the resistance has atemperature coefficient which counterbalances the temperaturecoefficient of the temperature sensing transistor in order to produce anapproximately zero slope of the temperature reading versus temperatureresponse within a portion of a desired temperature measurement range.

According to another embodiment of the invention, an RF transponder ischaracterized by a resistance, a temperature sensing circuit comprisinga temperature sensing transistor which exhibits a predictable change inits base-emitter voltage due to temperature, and transistors connectedfor mirroring a temperature-indicative current through the temperaturesensing transistor and through the resistance, an external measuringcapacitor having a capacitance value which is fixed and substantiallyindependent of temperature and pressure, a relaxation oscillator circuitwhich utilizes the external measuring capacitor to convert thetemperature-indicative current to an output signal, and a data capturecircuit for converting the output signal to a reading which isproportional to the temperature-indicative current. The value for thecapacitance of the external measuring capacitor is selected so that thereading plotted versus temperature has a monotonic slope for all valuesof temperature within a desired temperature measurement range. Thisembodiment of the invention may also include the use of a thermistor forthe resistance.

According to the invention, a method of scaling the output of atransponder is characterized by the transponder generating atemperature-indicative current, and utilizing an external measuringcapacitor to convert the temperature-indicative current to a readingwhich is proportional to the temperature-indicative current; selectingthe external measuring capacitor to have a capacitance value which isfixed and substantially independent of temperature and pressure; andselecting a fixed value for the capacitance of the external measuringcapacitor so that the reading plotted versus temperature has a monotonicslope for all values of temperature within a desired temperaturemeasurement range.

Other objects, aspects, features and advantages of the invention willbecome apparent from the description that follows.

BRIEF DESCRIPTION OF THE DRAWINGS

Reference will be made in detail to preferred embodiments of theinvention, examples of which are illustrated in the accompanyingdrawings. The drawings are intended to be illustrative, not limiting.Although the invention will be described in the context of thesepreferred embodiments, it should be understood that it is not intendedto limit the spirit and scope of the invention to these particularembodiments.

Certain elements in selected ones of the drawings may be illustratednot-to-scale, for illustrative clarity.

Often, similar elements throughout the drawings may be referred to bysimilar references numerals. For example, the element 199 in a figure(or embodiment) may be similar in many respects to the element 299 in another figure (or embodiment). Such a relationship, if any, betweensimilar elements in different figures or embodiments will becomeapparent throughout the specification, including, if applicable, in theclaims and abstract.

In some cases, similar elements may be referred to with similar numbersin a single drawing. For example, a plurality of elements 199 may bereferred to as 199 a, 199 b, 199 c, etc.

The cross-sectional views, if any, presented herein may be in the formof “slices”, or “near-sighted” cross-sectional views, omitting certainbackground lines which would otherwise be visible in a truecross-sectional view, for illustrative clarity.

The structure, operation, and advantages of the present preferredembodiment of the invention will become further apparent uponconsideration of the following description taken in conjunction with theaccompanying drawings, wherein:

FIG. 1 is a generalized diagram of an RF transponder system comprisingan external reader/interrogator and an RF transponder within a pneumatictire, according to the prior art;

FIG. 2 is a block diagram of major components of an RF transponder,according to a previous model of the invention;

FIG. 3 is a schematic diagram of major portions of the RF transponder ofFIG. 2, according to a previous model of the invention;

FIG. 3A is a schematic diagram of a portion of the RF transponder ofFIG. 2, according to a previous model of the invention;

FIG. 3B is a schematic diagram of a portion of the RF transponder ofFIG. 2, according to a previous model of the invention;

FIG. 3C is a diagram of a memory space within the RF transponder of FIG.2, illustrating how data may be arranged and transmitted, according to aprevious model of the invention;

FIG. 3D is a plot of transponder readings versus transponder power forthe RF transponder of FIG. 2, according to a previous model of theinvention;

FIG. 4A is a block diagram of major components of an RF transponder,according to the invention;

FIG. 4B is a diagram of a memory space within the RF transponder of FIG.4A, illustrating how data may be arranged and transmitted, according tothe invention;

FIG. 5 is a schematic diagram of a current scaling portion and arelaxation oscillator portion of the RF transponder of FIG. 4A,according to the invention;

FIG. 5A is a schematic diagram of a logic portion of the relaxationoscillator portion of FIG. 5, according to the invention;

FIG. 6 is a graph of temperature counts versus temperature for RFtransponders according to the invention wherein temperature sensingcircuit resistance is provided by a fixed resistor in one curve and by athermistor in other curves;

FIG. 7 is a schematic diagram of a temperature sensing circuit withexternal resistance provided by a thermistor, according to theinvention; and

FIG. 8 is a graph of N_(T) and N_(P) counts versus temperature for RFtransponders according to the invention wherein the pressure sensingmode is utilized to produce a temperature measurement response whichdoes not rollover like the temperature sensing mode response curve.

DETAILED DESCRIPTION OF THE INVENTION

It is an object of the present invention to provide a system formonitoring vehicle tire pressure and warning the driver when a low tireinflation pressure condition occurs.

FIG. 1 illustrates an RF transponder system 100 of the prior art,comprising an RF (radio frequency) transponder 102 disposed within(e.g., mounted to an inner surface of) a pneumatic tire 104. (Anantenna, not shown, is mounted within the tire 104 and is connected tothe transponder 102.) The transponder 102 is an electronic device,capable of transmitting an RF signal comprising unique identification(ID) information (e.g., its own serial number, or an identifying numberof the object with which it is associated -in this example, the tire104) as well as data indicative of a parameter measurement such asambient pressure sensed by a sensor (not shown) associated with thetransponder 102 to an external reader/interrogator 106. The externalreader/interrogator 106 provides an RF signal for interrogating thetransponder 102, and includes a wand 108 having an antenna 110, adisplay panel 112 for displaying information transmitted by/from thetransponder 102, and controls (switches, buttons, knobs, etc.) 114 for auser to manipulate the functions of the reader/interrogator 106.Although shown as a hand-held device, the reader/interrogator may be anelectronic unit mounted in a vehicle (not shown). The present inventionis directed primarily to implementing an RF transponder.

As is known, the ID and/or parameter measurement information may beencoded (impressed) in a variety of ways on the signal transmitted bythe transponder 102 to the reader/interrogator 106, and subsequently“de-coded” (retrieved) in the reader/interrogator 106 for display to theuser. The RF transponder 102 may be “passive”, in that it is powered byan RF signal generated by the external reader/interrogator 106 andemitted by the antenna 108. Alternatively, the RF transponder 102 may be“active”, in that it is battery-powered. Transponder systems such as thetransponder system 100 described herein are well known.

Commonly-owned, copending PCT Patent Application No. PCT/US98/07338filed Apr. 14, 1998 by Pollack, Brown, Black, and Yones (status:pending), incorporated in its entirety by reference herein, discloses atransponder, particularly a “passive” transponder which derives itsoperating power from an external radio frequency (RF) source, and whichis associated with a pneumatic tire for use in tire identification andtransmission of pressure and/or temperature data.

The aforementioned patent application PCT/US98/07338 discloses atransponder which is a previous model (model number “3070C”) of thetransponder of the present invention. Since the present inventioninvolves commonalities with, as well as improvements upon the previousmodel, relevant portions of the previous model will be describedhereinbelow, with reference to FIGS. 2, 3, 3A, 3B and 3C.

FIG. 2 is a block diagram of the model 3070C RF transponder 200 (compare102), illustrating the major functional components thereof. Thisexemplary system is described as an embodiment which preferably measurespressure and temperature, but it is within the scope of the invention toinclude measurement of other parameters which employ suitable sensors.

The transponder 200 is preferably implemented on a single integratedcircuit (IC) chip shown within the dashed line 202, to which areconnected a number of external components. Other dashed lines in thefigure indicate major functional “blocks” of the transponder 200, andinclude a transponder “core” 204 and a sensor interface 206. Thecomponents external to the IC chip 202 include an antenna system 210comprising an antenna 212 and typically a capacitor 214 connected acrossthe antenna 212 to form an L-C resonant tank circuit, an externalprecision resistor (Rext) 216, an external pressure-sensing capacitor(C_(P)) 218, and an optional external maximum temperature measurementswitch (MTMS) 220. The antenna 212 may be in the form of a coil antenna,a loop antenna, a dipole antenna, and the like. Alternatively, thesignal output by the transponder may be provided on a transmission line.For some of these antenna embodiments (e.g., a loop antenna), thecapacitor 214 may be omitted since it would not be of benefit in tuningsuch an antenna system. In the main hereinafter, a transponder having acoil antenna is described.

The pressure-sensing capacitor C_(p) is preferably a rugged, lowtemperature coefficient, sensor with a capacitance versus pressureresponse having good sensitivity and linearity in the pressure range ofinterest. An example is an all-silicon “touch mode” capacitive pressuresensor such as are known in the art, and mentioned hereinabove.

The transponder core 204 includes interface circuitry 222 for processingan RF signal, such as a 125 kHz (kilohertz) un-modulated carrier signalreceived by the antenna 212, for rectifying the received RF signal, andfor providing voltages for powering other circuits on the IC chip 202.For example, the interface circuitry provides a regulated supply voltage(Vdd) of 2.5 volts, and a temperature-independent bandgap voltage (Vbg)of 1.32 volts. The provision of various supply and reference voltagesfor the transponder circuitry are described in greater detailhereinbelow, with reference to FIG. 3B. The interface circuitry 222 alsoprovides the received RF signal, preferably at the input frequency (Fi)it is received, to a clock generator circuit 224 which generates clocksignals in a known manner for controlling the timing of other circuitson the IC chip 202, as well as the output frequency (Fc) of a signalwhich is transmitted by the transponder 200 to the externalreader/interrogator (e.g., 106).

A timing generator/sequencer circuit 226 receives the clock pulses fromthe clock generator circuit 224 and processes (e.g., divides) the clockpulses to generate timing windows (W_(T) and W_(P), describedhereinbelow) for predetermined periods of time (t_(T) and t_(P),respectively) during which parameter (e.g., temperature and pressure)measurements are made. The timing windows W_(T) and W_(P) may either beof substantially equal duration or of unequal duration. The timinggenerator/sequencer circuit 226 also controls the timing and sequence ofvarious functions (e.g., pressure measurement and capture, temperaturemeasurement and capture, described in greater detail hereinbelow)performed in the sensor interface 206, and is preferably implemented asan algorithmic state machine (ASM).

The transponder core 204 further includes a register/counter circuit 230which includes a temperature register 232 (e.g., 12-bit) and a pressureregister 234 (e.g., 12-bit) for capturing and storing temperature andpressure measurements (counts), respectively, and a block 236 ofaddressable memory (e.g., 120-bit), which includes an EEPROM array. Theregisters 232 and 234 and EEPROM array 236 are shown in a dashed line238 representing a block of addressable memory on the IC chip 202.

The register/counter circuit 230 also includes a multiplexer and columndecoder 240, as well as a row decoder 242 for controlling the sequencein which signals (i.e., data) are output on a line 244 to a modulationcircuit 246 which, via the interface circuitry 222, communicatesselected measured tire operating characteristics in a data stream viathe antenna system 210 to an external reader/interrogator (e.g., 106).

The transponder core 204 also includes a baud rate generator 248 whichcontrols the rate at which modulating information (e.g., the temperatureor pressure measurement) is applied to the modulation circuit 246. Thebaud rate generator 248 also provides a data carrier clock controllingthe output frequency Fc of the transponder and a data rate clockcontrolling a rate at which the data stream including measurements,calibration information, identification, etc. is modulated onto thetransponder 200 output carrier signal.

The sensor interface 206 includes a circuit 250 for generating an outputcurrent I(T)/N on a line 251 which is related to a predictablecharacteristic voltage of a temperature-sensitive component (e.g., Vbeof a transistor Q1, described hereinbelow) which is superimposed on theexternal resistor (Rext) 216. The output current I(T)/N on the line 251is provided to a relaxation oscillator 252. In general terms, therelaxation oscillator 252 oscillates at a frequency controlled by a rateof voltage change (dV/dT) which is a function of the output currentI(T)/N on line 251 and of internal capacitances C_(FX1), C_(FX2)associated with the relaxation oscillator 252 as well as an externalcapcitance (C_(P)) 218 that can be switched into the oscillator circuit.An output signal Fosc′ from the relaxation oscillator 252 is provided ona line 253 which, as will be explained in greater detail hereinbelow, isindicative of both ambient temperature and ambient pressure. As usedherein, the term “ambient” refers to the parameter being measured in thevicinity of the transponder 200, or more particularly in the vicinity ofthe respective sensors associated with the transponder 200. When thetransponder 200, 102 is mounted within a pneumatic tire (e.g., 104),“ambient pressure” and “ambient temperature” refer to the pressure andtemperature of the inflation medium (e.g., air) within the tire 104.

In operation, an RF signal from an external source (i.e.,reader/interrogator, not shown, compare 106) is received by the antenna212. This RF signal is rectified and used to power the RF transponder200. Modulating information applied to the modulation circuit 246 isused to alter characteristics of the antenna system 210 (e.g.,impedance, resonant frequency, etc.). These alterations are sensed bythe external reader/interrogator 106 and are decoded, providingcommunication of temperature and pressure information back from the RFtransponder 200 to the external reader/interrogator 106.

The timing generator/sequencer circuit 226 controls when the externalpressure-sensing capacitance (C_(P)) 218 is included in the generationof a signal at frequency Fosc′ which is output by the relaxationoscillator 252, and also controls the capturing of the pressure andtemperature counts via the data capture circuit 254. For example, tomeasure temperature, the temperature-sensitive current I(T) passesthrough the internal oscillator capacitors (C_(FX1) and C_(FX2)), butthe pressure-sensing capacitor (C_(P)) 218 is disconnected from (notincluded in) those capacitances. This means that the frequency Fosc′ ofthe oscillator output signal seen on line 253 is a function oftemperature alone. When the pressuresensing capacitor (C_(P)) 218 is“switched in”, then the output frequency Fosc′ of the oscillator 252 onthe line 253 will, as explained in greater detail hereinbelow, be afunction of both pressure and temperature. As described in greaterdetail hereinbelow, an algorithm is employed in the reader/interrogator106 to extract a “pressure-only” reading from the pressure-temperaturemeasurement.

It should be noted that references made herein to “pressure readings”,“pressure counts”, “pressure response”, “pressure register” and the likegenerally refer to “pressure” as measured by this transponder techniquewhich actually produces a hybrid pressure-temperature reading. When thishybrid reading has been processed to remove its temperature component,the reading will be referred to as a “pressure-only” reading.

As controlled by the timing generator/sequencer circuit 226, the datacapture circuit 254 directs the relaxation oscillator output signalFosc′ either to the temperature register 232 via line 255 or to thepressure register 234 via line 257, depending upon whether temperatureor pressure is being measured. Counters convert the oscillator frequencyFosc′ into counts which are stored in the registers 232, 234. The timing“window” provided by the timing generator/sequencer circuit 226 has aknown, controlled duration. As a result, the count remaining in(captured by) the respective temperature or pressure register (232, 234respectively) when the timing window “closes” is a function of(proportional to) the oscillation frequency Fosc′ of the relaxationoscillator 252, and therefore a function of temperature or pressure,whichever is being measured during that timing window.

The EEPROM array 236 is used to hold calibration constants that thereader/interrogator (e.g., 106) uses to convert temperature and pressurecounts (N_(T) and N_(P), respectively, described in greater detailhereinbelow) into temperature and pressure readings which can bedisplayed (e.g., via display 112) to a user. The EEPROM array 236 canalso store the ID of the transponder, calibration data for thetransponder, and other data particular to the given transponder.

FIG. 3 is a more-detailed schematic diagram 300 of several of thecomponents of the transponder 200 of FIG. 2, primarily those componentsdescribed hereinabove with respect to the sensor interface section 206of FIG. 2.

In this schematic diagram 300, conventional circuit symbols areemployed. For example, lines which cross over one another are notconnected to one another, unless there is a “dot” at their junction(cross-over), in which case the lines are connected with one another.Conventional symbols are employed for transistors, diodes, groundconnections, resistors, capacitors, switches, comparators, inverters,and logic gates (e.g., “AND”, “NAND”, “OR”, “NOR”).

The circuit is described in terms of a CMOS embodiment, wherein “P”followed by a number (e.g., “P1”) indicates a PMOS (P-channel)transistor and “N” followed by a number (e.g., “N1”) indicates an NMOS(N-channel) transistor. CMOS transistors are of the FET (field effecttransistor) type, each having three “nodes” or “terminals”—namely, a“source” (S), a “drain” (D), and a “gate” (G) controlling the flow ofcurrent between the source and the drain. In the description thatfollows, it will be evident that a number of the PMOS and NMOStransistors are “diode-connected”, meaning that their drain (D) isconnected to their gate (G). The general theory of operation oftransistors, particularly CMOS transistors, is well-known to thosehaving ordinary skill in the art to which the present invention mostnearly pertains.

As will be evident from the description that follows, a number of theCMOS transistors are connected in a “current-mirroring” configuration.The concept of current-mirroring is well known, and in its simplest formcomprises two similar polarity transistors (e.g., two PMOS transistors)having their gates connected with one another, and one of the pair oftransistors being diode-connected. Current-mirroring generally involvescausing a current to flow through the diode-connected transistor, whichresults in a gate voltage on the diode-connected transistor required toproduce that current. Generally, the gate voltage of the diode-connectedtransistor is forced to become whatever voltage is necessary to producethe mirrored current through that transistor. Since the diode-connectedtransistor, by definition, has no gate current, by applying the gatevoltage of the diode-connected transistor to any otheridentically-connected transistor, a mirrored-current will flow throughthe identically-connected transistor. Typically, the current-mirroringtransistors all have the same physical area, in which case the mirroredcurrent will be essentially the same as the current which is beingmirrored. It is also known to produce a mirrored current which is eithergreater than or less than the current being mirrored by making one ofthe transistors physically larger or smaller (in area) than the other.When such identically-connected transistors having different areas areconnected in a current-mirroring configuration, their scaled (larger orsmaller) areas will produce correspondingly scaled (larger or smaller)currents.

In the main hereinafter, the numerous connections between the variouscomponents of the circuit are clearly illustrated in the figure, and thedescriptive emphasis is on the various functions of and interactionsbetween the various components of the circuit rather than on reciting(ad nauseam) each and every individual connection between the variouscomponents, all of which are explicitly illustrated in the figure.

The antenna system 210 comprises a coil antenna 212 and an optionalcapacitor 214 (connected across the antenna 212 to form an L-C resonanttank circuit) providing an alternating current (AC) output to afull-wave rectifier circuit 302.

The full-wave rectifier circuit 302 (compare 222) comprises two PMOStransistors and two diodes, connected in a conventional manner, asshown, and outputs a full wave rectified direct current (DC) voltage ona line 303. A capacitor 304 is connected between the line 303 and groundto “smooth out” (filter) variations (“ripple”) in the full waverectified DC voltage on the line 303. The voltage on the line 303 thusbecomes a usable voltage for the remaining components of thetransponder—in this case, a positive supply voltage Vcc on the line 303.

A temperature-sensing circuit 306, corresponding approximately to thebase-emitter voltage-to-current converter 250 of FIG. 2, is connectedbetween the line 303 (Vcc) and ground, and includes four CMOStransistors labeled P1, P2, N1 and N2 and a lateral bipolar transistorlabeled Q1, and is connected to the external resistor 216 (Rext). Thetransistors P2 and N1 are diode-connected, as illustrated. The twotransistors P1 and P2 are connected in a current-mirroringconfiguration, and the two transistors N1 and N2 are also connected inwhat can generally be considered to be a current-mirroringconfiguration. The source (S) of the transistor N1 is connected via thetransistor Q1 to ground, and the source of the transistor N2 isconnected via the external resistor (Rext) 216 to ground.

As will become evident, the ability of the temperature-sensing circuit306 to produce a signal (i.e., a current) that is proportional to asensed ambient temperature (e.g., within the tire with which thetransponder is associated) is largely dependent on the characteristicthat the base-emitter voltage of the transistor Q1 is a highlypredictable and repeatable function of temperature. The resistor (Rext)216 is an external, precision, reference resistor, whose value issubstantially independent of temperature (as contrasted with thetemperature dependency of the transistor Q1). A suitable value for theresistor (Rext) 216 is, for example, 20.5 kilohms or 455 kilohms.

The transistor N2 is connected between the transistor P2 and theexternal resistor 216 (Rext) in a “source-follower” mode. As a voltageis impressed on the gate (G) of the transistor N2, its source voltagewill “follow” its gate voltage (minus an inherent voltage drop (Vgs)between its gate and its source).

As current flows through the transistor N1, its gate voltage will beoffset by its gatesource voltage drop (Vgs) above the emitter voltage atthe transistor Q1. Since the transistors N1 and N2 are essentiallyidentical, with the same current flowing through each of the twotransistors N1 and N2, they will have identical gate-source voltagedrops (Vgs). As a result, the voltage at the source of the transistor N2across the external resistor 216 (Rext) will be essentially identical tothe voltage at the emitter of the transistor Q1. Hence, applying Ohm'slaw (E=IR, or I=E/R), the current through the external resistor 216(Rext) will equal the emitter voltage of the transistor Q1 divided bythe resistance of the external resistor 216 (Rext).

In normal operation, all of the current flowing through the externalresistor (Rext) 216 flows through the source of the transistor N2 and,consequently, through the diodeconnected transistor P2. By acurrent-mirroring connection, the current through the transistor P2 isreplicated (mirrored) in the transistor P1. This ensures that thecurrent flowing through the transistors N1 and N2 will be the same, atall times, which further helps to ensure that the emitter voltage at thetransistor Q1 and the voltage across the external resistor (Rext) 216are identical, independent of voltage and process variations. Asmentioned hereinabove, the transistors N1 and N2 are connected in whatcan generally be considered to be a current-mirroring configuration.However, since they are not strictly identically connected, theirfunction in the circuit 306 is principally for “matching” Q1 and Rext.

In essence, the circuit 306 ensures that the current I(T) flowingthrough the external resistor (Rext) is predictable, and is a functionof the absolute temperature (T) of the transistor Q1. As described ingreater detail hereinbelow, this temperature-dependent current I(T)flowing through the external resistor (Rext) 216 is mirrored to arelaxation oscillator (312, described hereinbelow) to provide a signalindicative of the temperature of the transistor Q1 to the externalreader (106, FIG. 1). As described in greater detail hereinbelow, theoutput frequency Fosc′ of the relaxation oscillator 312 will be afunction of the absolute temperature (T) of the transistor Q1.

At this point, it is useful to note that it is essentially thetransistor Q1 that is being employed as the temperature-sensing elementof the overall transponder circuit. The transponder circuitadvantageously employs an inherent characteristic of such a transistorimplemented in CMOS technology that the base-emitter voltage of thetransistor Q1 will vary by a predictable amount of −2.2 mv/° C.(millivolts per degree Celsius).

It should be noted that the transponder of the present invention isdescribed in terms of a “passive” device, relying on RF energy beingsupplied to it by an external source (106, FIG. 1) to power up itscircuitry. However, it is within the scope of this invention that thetransponder contains its own power supply, such as in the form of abattery. In either case, when first powering up circuitry such asdescribed with respect to the temperature-sensing circuit 306, it isimportant to ensure that they “ramp up” to their normal operating statefrom their quiescent state in a reliable and predictable (controlled)manner. To this end, two lines 305 and 307 are illustrated which areconnected between the temperature-sensing circuit 306 and a “startup”circuit 308.

The startup circuit 308 (also part of the base-emittervoltage-to-current converter 250 of FIG. 2) is connected between thesupply voltage (Vcc) on the line 303 and ground, and serves two mainpurposes: (i) to get current flowing in the temperature-sensing circuit306 when the transponder (200) first starts up from a powered downstate; and (ii) to mirror and convert the current flowing through thetransistor P2 from a supply-referenced current to a ground-referencedcurrent.

Startup is initiated by the transistor P3. The transistor P3 isfabricated to have high channel resistance so as to function in a “weakpull-up” mode. With its gate connected to ground, it will always be“on”, and will behave essentially like a resistor having a highresistance (e.g., >1 mega-ohm).

Since, at startup, no current flows elsewhere in the circuit, thetransistor P3 operates to pull the gate of the transistor N3 towards thesupply voltage (Vcc), thereby turning the transistor N3 “on”, whicheffectively connects the grounded source of transistor N3 to its drain(D) which, in turn, grounds the gates of transistors P1, P2, and P4, andalso grounds the drain of diode-connected transistor P2. This causescurrent to flow through the transistor P2 of the temperature-sensingcircuit 306 into the drain of the transistor N3. Since the transistorsP1, P2 and P4 are current-mirror connected (via the “Pbias” line 305),the current now flowing through transistor P2 will be mirrored in thetransistors P1 and P4. As current flows through the transistor P4 intothe diode-connected transistor N5, a current-mirroring connectionbetween the transistors N4 and N5 causes a corresponding current to flowthrough the transistor N4, thereby pulling the gate of transistor N3 toground, thereby effectively shutting “off” the flow of current throughthe transistor N3.

However, with current now flowing through the current-mirroredtransistors P1, P2 and P4, current flowing from the transistor P1through the diode-connected transistor N1 into the transistor Q1 forcesthe temperature-sensing circuit 306 to “start up” in its stableoperating point state (rather than its zero current state). Afterstartup, the transistor N3 essentially “drops out” of the circuit,having performed its intended function.

The transistor N5 is connected in a current-mirroring configuration withthe transistor N4 (and, as described hereinbelow, with the transistorN6). Therefore, essentially, with a current equivalent to the currentthrough the external resistor (Rext) 216 flowing through the transistorN5, the same current flows through the transistor N4, therebyestablishing a reference voltage (Nbias) on the line 309. The referencevoltage (Nbias) on the line 309, as well as a supply voltage (Vdd) on aline 309′, are provided to a current scaling circuit 310.

The supply voltage (Vdd) on the line 309′ is provided in any suitablemanner, such as a multiple of a bandgap voltage (Vbg) generated in aconventional manner elsewhere on the chip, and its magnitude (e.g., 1.32volts) should be independent of temperature, such as inherent to thesilicon process which is employed in making the chip. The provision ofsuch a stable (e.g., bandgap) voltage (e.g., Vbg) and the supply voltage(e.g., Vdd) derived therefrom is well within the purview of one havingordinary skill in the art to which the present invention most nearlypertains, and is described in greater detail hereinbelow with respect toFIG. 3B.

The current scaling circuit 310 (also part of the base-emittervoltage-to-current converter 250 of FIG. 2) is constructed in thefollowing exemplary manner. The sources of the transistors P5 and P6 areconnected to supply voltage Vdd. The gate of a transistor N6 receivesthe reference voltage (Nbias) on the line 309. The transistor N6 isconnected in a current-mirroring configuration with the transistor N5(as well as with the aforementioned transistor N4) and will thereforemirror the flow of current I(T) through the transistors N4 and N5.Consequently, the flow of current through the diode-connected transistorP5 will mirror the flow of current through the transistors N4, N5 andN6.

The transistors P5 and P6 are connected in a current-mirroringconfiguration, but are fabricated (using conventional CMOS fabricationtechniques) such that current flowing through the transistor P6 isscaled up or down by a ratio (N) of the physical area of the transistorP5 to the physical area of the transistor P6. For example, if thetransistor P6 is smaller in size than the transistor P5 (i.e., thetransistor P5 is “N” times larger in area than the transistor P6), thenthe current flowing through the transistor P6 will be commensurately (Ntimes) smaller than the current flowing through the transistor P5. Thus,the “scaled” current flowing through the transistor P6, is labeled“I(T)/N” in the figure, and is provided on a line 311 (compare 251) to arelaxation oscillator circuit 312 (compare 252). It is well known thatthe ratio of the currents between the transistors P5 and P6 can readilybe established by conventional circuit processing techniques, such as bysimply making one of the transistors larger than the other, or byimplementing a one of the two transistors as the aggregate of two ormore same-size transistors so that their aggregate area is larger thanthe area of the other of the two transistors.

The relaxation oscillator circuit 312 (compare 252) is of fairlyconventional design, and includes a measurement switching circuit 315 atthe “front end” of a set-reset circuit 314 comprising two phase paths314 a, 314 b. This circuit 315 comprises a pair of complementarytransistors P7 and N7 connected to a charged side of a measurementcapacitor C_(FX1) at the front end of a one phase path (φ1) 314 a; andanother pair of complementary transistors P8 and N8 connected to acharged side of another measurement capacitor C_(FX2), plus a switch 350to add another measurement capacitor C_(P), all at the front end ofanother phase path (φ2) 314 b.

Connected as illustrated, for a given pair of transistors (e.g., P7 andN7), when their common gate voltage is high (i.e., towards positivesupply) their output (e.g., to phase path 314 a) will be grounded(connected to ground and isolated from current I(T)/N on line 311), andwhen their common gate voltage is low, their output will provide thecurrent I(T)/N flowing on the line 311 to a respective one of the phasepaths (e.g., 314 a) of the relaxation oscillator 312. As is known forcircuitry such as the relaxation oscillator 312, when the common gatevoltage of a one of the pairs of transistors (e.g., P7 and N7) is high,the common gate voltage of the other of the pairs of transistors (e.g.,P8 and N8) will be low, and vice-versa. In this manner, each phase path314 a and 314 b has a duty cycle (i.e., its “on” time), which may be thesame as or may be different than the duty cycle of the other phase path314 b and 314 a, respectively. Thus, each pair of transistors (e.g., P7and N7) may be considered to be an “input switch” to its respectivephase path (e.g., 314 a).

Each phase path 314 a and 314 b of the relaxation oscillator 312 has acomparator 316 a and 316 b, respectively, at its input, and has afixed-value capacitor C_(FX1) and C_(FX2), respectively, connectedbetween the negative (−) input of the comparators 316 a and 316 b andground. The capacitors C_(FX1) and C_(FX2) have exemplary capacitancevalues of 2-5 pf (picofarads) and 2-5 pf, respectively, and arepreferably implemented as equal-valued “onchip” devices, such aspoly-to-poly capacitors exhibiting a low temperature coefficient (e.g.,less than 20 ppm). The positive (+) inputs (terminals) of thecomparators 316 a and 316 b are tied together and are set to a referencethreshold voltage Vbg, such as 1.32 volts, which is independent oftemperature.

A “NOR” logic gate 318 a and 318 b is connected at the output of eachphase path 314 a and 314 b, respectively, and the two NOR gates 318 aand 318 b are cross-connected to form a latching circuit having anoutput on a line 319 a and 319 b. The cross-connected NOR gates 318 aand 318 b are thus capable of functioning as a flip flop, or an RS(re-set/set) latch.

When the common gate voltage of one of the input switches (e.g., P7 andN7) is high, the respective capacitor (e.g., C_(FX1)) for that phasepath (e.g., 314 a) is grounded (shorted out, caused to be devoid ofcharge). Conversely, when the common gate voltage of one of the inputswitches (e.g., P7 and N7) is low, the scaled current I(T)/N from line311 is applied to (allowed to flow into) the respective capacitor (e.g.,C_(FX1)) for that phase path (e.g., 314 a), and the capacitor begins tocharge (acquire an increasing voltage across the capacitor). When thevoltage across the capacitor C_(FX1)/C_(FX2) reaches the comparatorreference voltage Vbg the output of the comparator 316 a/316 b goes lowand changes the state of the output of the latch 318 a/318 b on the line319 a/319 b. In this manner, the relaxation oscillator 312 willoscillate at a frequency Fosc determined by the rise time of thecapacitors C_(FX1) and CFX₂ and, importantly, by the scaled currentI(T)/N being supplied to the capacitors C_(FX1) and CFX₂. With greatercurrent I(T)/N being supplied, the voltages of the capacitors C_(FX1)and CFX₂ will rise faster, crossing the threshold voltage faster, andcausing the relaxation oscillator 312 to oscillate faster, therebyincreasing the frequency Fosc of the signal on the line 319 a. Thesignal on the line 319 a is inverted by an inverter 320, as shown, toprovide a signal of frequency Fosc′ on the line 321.

As described in greater detail hereinbelow, the oscillator 312 iscontrolled to run in two mutually-exclusive modes, a temperature-sensingmode (between times t0 and t1) and a pressure-sensing mode (betweentimes t1 and t2), as controlled by the timing generator/sequencer 226.The frequency of the oscillator output signal Fosc (and Fosc′) will bedifferent in each of these two modes.

GENERATING TEMPERATURE AND PRESSURE READINGS

In the exemplary context of the transponder 200 being associated with apneumatic tire, it is principally desirable to determine the pressurewithin the pneumatic tire. For example, a typical passenger vehicle tiremay be properly inflated at about 32 psi (about 221 kPa). Since tireinflation pressures are normally specified as “cold” pressures (pressuremeasured when the tire is not heated by operation), and since amonitoring device will be reporting pressures measured in tires whichare most likely in use and therefore “hot”, it is secondarily desirableto determine the temperature of the inflation medium (e.g., air) withinthe pneumatic tire. Utilizing the temperature measurement, a monitoringsystem (e.g., 106) can, for example, convert the measured pressure to a“cold” pressure with simple calculations based on the ideal gas law(PV=μRT). This “cold” pressure could be considered a“temperature-independent” pressure, which is also an indication of themass of air contained by the tire. With reference to the transponder200, the hybrid “pressure” measurement it produces must be converted (bydifferent calculations detailed hereinbelow) to a true pressure-onlymeasurement before it can be used in such gas-law calculations.

It is, for example, estimated that an approximate 10% decrease in fuelconsumption could be realized if the pneumatic tires on vehicles wereoperated at their specified pressure.

Although vehicle fleet operators are typically sensitive to this issue,and check and adjust tire pressure frequently, the average operator of apassenger vehicle is often less inclined to keep an eye on their tirepressure until, for example, the tire is visibly flattened out. In suchcases, an LCD (liquid crystal display) readout or the like on thedashboard of a car could provide dynamic tire inflation information tothe operator of a vehicle, the pneumatic tires of which are equippedwith a transponder such as the one described herein. Of no lesssignificance is the emergence of “run-flat” tires being marketed byvarious tire manufacturers. The Goodyear EMT (extended mobilitytechnology) series of tires is an example of a “run-flat” tire, anoverall purpose of which is to allow a driver to travel up to 50 miles(approximately 120 kilometers) on a deflated (“flat”) tire, at“reasonable” operating speeds (e.g., 60 miles per hour, or 144kilometers per hour), while maintaining normal control over the vehicle.Such run-flat tires are generally well known, and do not form a portionof the present invention, per se. When running “flat” on a run-flattire, it is particularly important that the driver be alerted to thefact that he or she is operating the vehicle on “borrowed time” asindicated, principally, by an indication, whether visual or audio (e.g.,a beep) that the tire is indeed “flat” and needs to be repaired orreplaced at his or her earliest convenience (and before the run-flatmileage limit).

By allowing the relaxation oscillator 312 to run, the frequency of itsoutput signal Fosc (and Fosc′) will be a function of the absolutetemperature of (sensed by) the transistor Q1. This is true in both thetemperature-sensing mode and the pressure-sensing mode of operation.

In the temperature-sensing mode, and in the case that the capacitancevalues for C_(FX1) and C_(FX2) are equal, which is preferred, therelaxation oscillator 312 will have a symmetrical (balanced, 50%) dutycycle. In the pressure-sensing mode, the pressure-sensing capacitor(C_(P)) 218 is switched by a semiconductor switch 350 across C_(FX2),which changes the duty cycle and output frequency Fosc (and Fosc′) ofthe relaxation oscillator 312.

In the temperature-sensing mode, only the fixed capacitors C_(FX1) andC_(FX2) are being alternately charged (and discharged) resulting in a50% duty cycle with a period proportional to ambient temperature. In thepressure-sensing mode, the pressure-sensing capacitor (C_(P)) 218 isswitched into phase path 314 b of the oscillator 312. Thus, for a giventemperature, for the first half of the oscillator period the phase path314 a behaves in the same manner as in the temperature-sensing mode, andfor the second half of the oscillator period the phase path 314 bbehaves in a manner that is proportional to the capacitance value of thefixed capacitor C_(FX2) plus the capacitance value of thepressure-sensing capacitor (C_(P)) 218. This, in effect, slows down theoscillator and changes its duty cycle. The change in the duty cycle isindicative of the ratio of C_(P) to C_(FX2). Thus, from the ratio of thetwo periods (with and without C_(P) in the circuit, it isstraightforward to calculate what the additional capacitance C_(P) is,hence the sensed pressure. As described in greater detail hereinbelow,the temperature-dependency of the oscillator output in thepressure-sensing mode can be completely eliminated, in a straightforwardmanner.

The “slowing down” of the oscillator when the pressure-sensing capacitor(C_(P)) 218 is switched into the oscillator circuit results, inevitably,in there being relatively fewer oscillator output pulses (reduced outputfrequency) to count during a given pressuremeasurement window (e.g.,W_(P)) than during a similar duration temperature-measurement window(e.g., W_(T)). In other words, a “slowed-down” oscillator will reducethat rate at which counts indicative of the parameter measurement arecollected. In order to increase the resolution (quantity) of the counts(N_(P)) generated during the pressure-measurement window (W_(P)), it iscontemplated that the pressure-measurement window (W_(P)) can beincreased in size (changed in duration) so as to allow for the captureof an appropriate number of pressure counts in the pressure register234. This can readily be accomplished simply by establishing a larger(than otherwise) value for the time t2 which establishes the end of thepressure-measurement window (W_(P)) in the pressure-sensing mode(between times t1 and t2), as controlled by the timinggenerator/sequencer 226. For example, the temperature-measurement windowW_(T) (between times t0 and t1) can be on the order of several ones(e.g., eight) of milliseconds, and the pressure-measurement window W_(P)can be on the order of tens or dozens (e.g., eighty) of milliseconds.Alternatively, it is contemplated that the scaled current (I(T)/N)flowing out of the current scaling circuit 310 to the relaxationoscillator 312 could be increased during the pressure-measurement window(W_(P)) to increase the fundamental frequency of the relaxationoscillator 312, thereby increasing the overall resolution of thepressure count. This can readily be accomplished, for example in thecase of transistor P6 being smaller in size (area) than the transistorP5, simply by switching in a transistor P6′ (not shown) in lieu of thetransistor P6, the transistor P6′ having a larger area than thetransistor P6 so that the ratio of the areas of the transistors P5 andP6 is closer to unity (i.e., less scaled down) and the current to therelaxation oscillator 312, hence its counting rate, is increased. Suchswitching in of another transistor P6′ is readily effected with a switch(not shown) comparable to the aforementioned switch 350 which switchesin the pressure-sensing capacitor (C_(P)) 218. One having ordinary skillin the art to which the present invention most nearly pertains willreadily understand how to offset the “slowing down” of the oscillatorwhen the pressure-sensing capacitor (C_(p)) 218 is switched into theoscillator circuit, in light of the teachings presented herein.

OPTIMIZING PRESSURE-RESPONSIVENESS

Obtaining (and displaying) an accurate pressure reading being ofparamount importance when monitoring the pressure of a pneumatic tire,certain parameters of the transponder circuit may be established tomaximize its pressure-responsiveness and therefore improve the accuracyof the pressure reading displayed by the external reader/interrogator(e.g., 106).

As described hereinabove, the transponder responds to the changingcapacitance of the pressure sensor (C_(P)) 218 by changing the value ofa binary 12-bit word that is transmitted to the externalreader/interrogator 106. This binary word is the count of an oscillatorfrequency during a timing window W_(P) (between t1 and t2) establishedby the timing generator/sequencer 226. The pressure response cantherefore be described as the change in counts per unit change incapacitance of the pressure-sensing capacitor (C_(P)) 218.

Pressure-responsiveness (and resolution) of the transponder has beenfound to be dependent on a number of factors, each of which can beanalyzed. For example, it has been determined that:

(a) Increasing the pressure-measurement window W_(P) to make it largerthan the temperature-measurement window W_(t) will increase the pressurecount N_(p) (and not the temperature count N_(T)) for a given value ofthe pressure-sensing capacitor (C_(P)) 218, to make up for therelatively lower oscillator frequency which occurs during pressuremeasurement compared to temperature measurement (as detailedhereinabove).

(b) Increasing the scaled current I(T)/N to the oscillator 312 willproportionally increase the pressure count N_(P) for a given value ofthe pressure-sensing capacitor (C_(P)) 218.

(c) Decreasing the values for capacitor(s) C_(FX1) and/or C_(FX2) willproportionally increase the pressure count N_(P) for a given value ofthe pressure-sensing capacitor (C_(P)) 218.

(d) Increasing the scaled current I(T)/N to the oscillator willproportionally increase the pressure count N_(P) (for a given value ofC_(P)) at a greater rate than decreasing the values for capacitorsC_(FX1) and C_(FX2).

(e) Increasing the scaled current I(T)/N will increase both the pressurecounts N_(P) and the temperature counts N_(T) unless the currentincrease can be made to occur only during the pressure-measurementwindow W_(P).

(f) Decreasing the values for capacitor(s) C_(FX1) and/or C_(FX2) willincrease both the pressure counts N_(P) and the temperature counts N_(T)even if only one of the capacitors is changed.

As a general proposition, increasing the pressure counts N_(P) isdesirable. However, one having ordinary skill in the art to which thepresent invention most nearly pertains will readily appreciate thatthere is a practical upper limit to increasing the pressure counts at afrequency which may become unacceptably large for the capability ofcertain circuits of the IC chip.

MEASURING PARAMETERS

FIG. 3A illustrates the components involved in the final step ofcapturing temperature and pressure measurements in the transponder. Thesignal Fosc′ output by the relaxation oscillator 312 is provided on line321 (compare 253) to an input of each of two AND gates 360 and 362 inthe data capture circuit 254. A signal (“Capture Temp”) is provided bythe timing generator/sequencer 226 to the other input of the AND gate360 during the temperature-sensing window (W_(T)) so as to load thetemperature register 232 via line 255 with the count (“data,” or“reading”) N_(T) indicative of measured temperature. Another data signal(“Capture Press”) is provided by the timing generator/sequencer 226 tothe other input of the AND gate 362 during the pressure-sensing window(W_(P)) so as to load the pressure register 234 with the count (“data,”or “reading”) N_(P) indicative of measured pressure. Each of theregisters 232, 234 has a counter (not shown) associated with it toconvert the incoming oscillating signal Fosc′ to a stored count. The twocounts N_(T), N_(P) are then shifted out of the registers 232 and 234,via the MUX 240, to the modulation circuit 246 described hereinabove.

When the transponder is powered up, temperature and pressure arecontinuously measured, and these measurements are transmitted back tothe external reader/interrogator 106 as data words in a data stream. Forexample, each of the temperature and pressure parameters can betransmitted back to the reader/interrogator 106 as 12-bit data words inselected (known) portions of a larger (e.g., 144-bit) data stream. Onebit in the overall data stream may be dedicated to the state (e.g.,“closed” or “open”) of the MTMS switch 220. A complete description of anexemplary data stream being transmitted by the transponder to theexternal reader/interrogator is set forth hereinbelow with reference toFIG. 3C.

Temperature is suitably measured by counting the number of cycles outputfrom the oscillator 312 during a fixed time period (window W_(T) of timefrom t0 to t1) having a time period t_(T). For example, a down-counter(not shown, but associated with the temperature register 232) may beclocked by the oscillator, such that at the end of the window W_(T) timeperiod t_(T), a temperature count N_(T) is generated. The relationshipbetween temperature count N_(T) and temperature is substantially linearfor the circuitry 300 of this embodiment.

Similarly, pressure can be measured by counting the number of cyclesoutput from the oscillator 312 during a fixed time period (window W_(P)of time from t1 to t2) having a time period t_(P). For example, adown-counter (not shown, but associated with the pressure register 234)may be clocked by the oscillator, such that at the end of the windowW_(P) time period t_(P), a temperature count N_(P) is generated. Therelationship between pressure count N_(P) and pressure is a predictablefunction of both actual pressure and temperature for the circuitry 300of this embodiment. As explained hereinbelow, by manipulating thetemperature and “pressure” counts (N_(T) and N_(P)) this hybridpressure-temperature value can be used to determine a pressure-onlyvalue.

OBTAINING A PRESSURE-ONLY READING AT THE READER/INTERROGATOR

The fundamental frequency of the oscillator 312 is set by parameters inthe IC chip (e.g., 202) and, as described hereinabove, istemperature-dependent. Therefore, the pressure response N_(P) is afunction (hybrid) of both temperature and pressure, and the relationshipof N_(P) to C_(P) is nonlinear. Therefore, using a linear equation forcalculating the pressure response would inevitably lead to significanterrors over a range of pressures being measured. However, for limitedranges of pressures being measured, for example over a 20 psi (138 kPa)range of pressures, using a linear equation may be acceptable. A betterapproximation might be obtained using a polynomial equation, but thiswould complicate the reader/interrogator logic, making for slowerresponse, and would require additional calibration constants.

An important advantage of using the transponder circuitry describedhereinabove is that the relationship of N_(T)/N_(P) to pressure sensorcapacitance C_(P) is linear, and requires no temperature compensationterm in the equation (algorithm) used by the reader/interrogator 106 tocalculate pressure, thereby greatly simplifying the design of thereader/interrogator. (This also assumes the use of a pressure sensor(C_(p)) 218 which has a substantially linear relationship betweenpressure and capacitance.) This beneficial “ratiometric” relationship isreadily demonstrated by the following equations:

Generally,

count=counting window time (t)*frequency (F)

F=1/period

Charging time=V*C/I

for a capacitor with capacitance C to be charged to a voltage V with acurrent I.

Since the period of the relaxation oscillator 312 with output signal offrequency Fosc′ is the sum of the charging times for the capacitances inthe two phase paths 314 a and 314 b, the above equations can bemanipulated to obtain a general equation for the count from such arelaxation oscillator with capacitances C_(FX1) and C_(FX2), forexample:

count=t/(V*C_(FX1)/I+V*C_(FX2)/I)=t*I/(V*(C_(FX1)+C_(FX2)))

Substituting the values for the temperature and pressure counts:

N_(T)=(t_(T)*I(T)/n_(T))/(Vbg*(C_(FX1)+C_(FX2)))  [EQ. A]

N_(P)=(t_(P)*I(T)/n_(P))/(Vbg*(C_(FX1)+C_(FX2)+C_(P)))

where n_(T) and n_(P) are values for the scaling factor N in the scaledcurrent I(T)/N which could be different during the temperature andpressure measurement windows, respectively.

Dividing equations to obtain N_(T)/N_(P):

N_(T)/N_(P)=(t_(T)/t_(P))*(n_(P)/n_(T))*(C_(FX1)+C_(FX2)+C_(P))/(C_(FX1) +C_(FX2))

or

N_(T)/N_(P)=(t_(T)/t_(P))*(n_(P)/n_(T))*(1+(C_(P)/(C_(FX1)+C_(FX2)))  [EQ.B]

Since everything to the right of the equals sign is a defined constantexcept for the pressuresensing capacitance C_(P), it can be seen thatthere is a linear relationship between N_(T)/N_(P) and C_(P) (and thuspressure). This means that N_(T)/N_(P) is only a function of pressure,and is insensitive to temperature or capacitor-charging currentvariations.

If none of the response optimization steps described hereinabove havebeen utilized, then the equation EQ. B can be simplified sincecapacitors C_(FX1) and C_(FX2) have the same value C_(FX); themeasurement windows W_(T) and W_(P) have the same time widtht_(T)=t_(P)=t_(W) (e.g., 8.192 ms); and the current scaling factorsn_(T) and n_(P) have the same value N:

 N_(T)/N_(P)=1+(C_(P)/2*C_(FX))

It can be seen from equation EQ. A that there is already a linearrelationship between the temperature count N_(T) and the current I(T)(which is, in turn, proportional to temperature).

In both of the measurement equations EQ. A and EQ. B it can be seen thatlinear relationships exist, but the slope and intercept of theseequations are complex combinations of multiple parameters which areunique to a given transponder design, and are likely to be differenteven for each transponder of a given design due to manufacturingvariances. In a simple embodiment of this invention, the transpondercould transmit only the counts N_(T) and N_(P) to a reader/interrogator,and the reader/interrogator would have to use assumed average values forslope and intercept in order to determine temperature and pressure. Thiscould cause significant inaccuracy, so the preferred embodiment asdescribed herein stores calibration constants in the transponder memory(e.g., 236) and transmits these calibration constants with themeasurement counts N_(T) and N_(P) so that the reader/interrogator(e.g., 106) can accurately calculate temperature and pressure usinglinear equations customized/optimized for the individual transpondergenerating the measurements. The linear equations used in the exemplaryreader/interrogator (e.g., 106) are of a well-known “point-slope” form:

y−y₁=m(x−x₁)

where:

(x₁,y₁) is the defining point; and

m is the slope.

The slope (m) can be determined from any two points on the line:(x₁,y₁), (x₂,y₂):

m=(y₂−y₁)/(x₂−x₁)

Substituting for x and y, a specific equation for the temperatureresponse line becomes:

N_(T)−N_(T1)=m_(T)(T−T₁)

Choosing a value such as 25° C. for temperature T₁ yields the equation:

N_(T)−N_(T25)=m_(T)(T−25)

Solving for N_(T) yields the following equation for the temperatureresponse line:

N_(T)=m_(T)(T−25)+N_(T25)

wherein the slope m_(T) of the temperature response line is:

m_(T)=(N_(T2)−N_(T1))/(T₂−T₁)

As long as the reader/interrogator “knows” the assumed defining pointtemperature (e.g., 25° C.), then it will be able to calculate atemperature (T) from a received value of temperature count N_(T) usingthe calibration constants: a defining point temperature count N_(T25)and a slope m_(T) which are also transmitted to the reader/interrogator.A similar set of equations can be applied to determine pressure fromtransmitted pressure (and temperature) counts and pressure calibrationconstants. As noted hereinabove, the pressure-only reading is bestdetermined from a linear equation utilizing a ratio N_(T)/N_(P)(temperature count divided by pressure count) instead of just thepressure count N_(P).

The calibration constants are determined in a calibration process whichincludes exposing each transponder to a set of controlled, knowntemperature and pressure conditions and recording the corresponding setof temperature and pressure counts (N_(T) and N_(P)) generated by thattransponder. Calculations on these calibration test results determinefour calibration constants which are then stored in the transpondermemory (e.g., 236). The four calibration constants are numbersrepresentative of the slope and defining point for a linear response oftemperature versus temperature count N_(T), and pressure (only) versuscount ratio N_(T)/N_(P).

GENERATING RELIABLE SUPPLY AND REFERENCE VOLTAGES

As described hereinabove, the positive (+) inputs (terminals) of thecomparators 316 a and 316 b are tied together and are set to a reference“bandgap” voltage Vbg, such as 1.32 volts, which is independent oftemperature. As also mentioned hereinabove, the supply voltage (Vdd) onthe line 309′ may be provided as a multiple of the reference bandgapvoltage (Vbg) so as to be a stable operating voltage for the currentscaling circuit 310 and the relaxation oscillator 312.

FIG. 3B illustrates a circuit 370 suitable for generating the supplyvoltage Vdd. A temperature-independent calculable bandgap voltage Vbg isreadily derived, based on the processing techniques employed infabricating the IC chip, as being inherent to the selected process(e.g., CMOS). This bandgap voltage Vbg is provided to the positive (+)input of an operational amplifier 372, connected as shown, in a feedbackloop having gain, to provide supply voltage Vdd as an integral multipleof the bandgap voltage Vbg.

AN EXEMPLARY DATA STREAM

As mentioned hereinabove, information (data) from the transponder istransmitted to the external reader/interrogator in the form of a datastream, a portion of which is the temperature count N_(T), anotherportion of which is the pressure count N_(P), and another portion ofwhich represents the state (e.g., “closed” or “open”) of the MTMS switch(220).

Remaining portions of the data stream may contain information which ispersonalized to a given transponder unit such as its ID information(e.g., serial number), calibration constants, and the like.

FIG. 3C illustrates an exemplary architecture for information which isstored in memory (e.g., 238) within the transponder 200, as well as adata stream which is transmitted by the transponder 200 to the externalreader/interrogator 106. The memory 238 of the transponder core 204 has,for example, a 144-bit address space which includes 119 (one hundrednineteen) bits of programmable memory and one address location dedicatedto the state of the MTMS switch 220—these 120 (one hundred twenty) bitsof programmable memory constituting the EEPROM 136—plus two 12-bittemperature and pressure registers 232 and 234, respectively.

Each of the 119 programmable memory bits can separately be written towith any combination of data, including synchronization (sync) patterninformation, general data, error checking codes, and temperature andpressure calibration data. The EEPROM is ‘block writeable’, meaning thatin the ‘write’ mode, the entire 120 bits of EEPROM are programmed to alogical (binary) value of “1”. Individual bits can be ‘erased’ (set to alogical value of “0” simply by clocking the chip to the bit's physicaladdress and placing the chip into the ‘erase’mode). The address locationis preserved.

In this example, the first twelve data locations (000 . . . 011 in ROW1) are reserved for sync. The next seventy one data locations (012 . . .082 in ROWs 2 through 7) are for general information and a value for adata validation algorithm such as CRC (Cyclic Redundancy Check). Thenext data location (083) contains the logic level (state) of the MTMSswitch 220. A logical value of “1” indicates that the MTMS switch isopen and a logical value of “0” indicates that the MTMS switch isclosed.

As mentioned hereinabove, each transponder unit is suitably calibratedprior to its installation in a tire. The next twelve data locations (084. . . 095 in ROW 8) hold temperature calibration (e.g., defining point)data (“TEMP COMP”). The next twelve data locations (096 . . . 107 in ROW9) hold pressure calibration (e.g., defining point) data (“PRESS COMP”).The next twelve data locations (108 . . . 113 and 114 . . . 119 in ROW10) hold calibration (e.g., slope) information for temperature andpressure, respectively.

As counts N_(T) and N_(P) for temperature and pressure are generated, asdescribed hereinabove, they are stored in ROWs 11 and 12 of the overallmemory space, which correspond to the temperature and pressure registers232 and 234, respectively. Various predetermined values can be stored toindicate error conditions such as overflow and short-circuit.

OPERATING FREQUENCIES AND MODULATION

The transponder of the present invention is not limited to anyparticular operating frequency. The choice of operating frequency willdepend largely upon factors such as where the transponder is mounted inrelationship to the object it is monitoring, the location of thereader/interrogator antenna (108), and relevant government regulationspermitting (conversely, restricting) data transmissions of the type setforth herein in selected portions of the overall RF frequency spectrum.

An example of suitable operating frequencies for operating thetransponder in the United States is 60 KHz to 490 KHz.

The transponder can be polled (and powered) by the reader/interrogator106 at a first “interrogation” frequency (Fi), and the data stream canbe transmitted back to the reader/interrogator at a second “datacarrier” frequency (Fc) which is, conveniently, a whole number multipleor fraction of the interrogating frequency. For example, Fc=Fi/2. Or,Fc=Fi/4. The frequency (Fc) at which the data stream is transmitted backto the reader/interrogator is independent of the data rate, which isestablished by the clock generator 224 and the baud rate generator 248.However, one having ordinary skill in the art to which the presentinvention most nearly pertains will recognize that the range ofavailable baud rates will typically be significantly less than theinterrogation frequency (Fi). The baud rate is preferably derived fromthe interrogation frequency (Fi) of the reader/interrogator, such as awhole number fraction thereof. For example, the baud rate may be set atFi/32 (or, in the case of Fc=Fi/2, the baud rate can be set to Fc/16).

For example, the interrogation frequency (Fi) may be 125 KHz, and thedata carrier (Fc) may be set to 62.5 KHz, or half of the interrogationfrequency.

In another example, an interrogation frequency (Fi) of 13.56 MHz hasbeen found to be suitable.

The data stream, such as the exemplary data stream described withrespect to FIG. 3C is impressed by the modulator circuit 246 onto theantenna 212, and transmitted to the reader/interrogator 106. It iswithin the scope of this invention that any suitable modulation schemebe employed, including amplitude modulation (AM), frequency modulation(FM), frequency shift keying (FSK), and phase shift keying (PSK).However, phase shift keyed (PSK) is preferred. AM modulation is notparticularly well-suited to digital transmission. Frequency modulationschemes such as FM or FSK may be somewhat problematic with regard topropagating the data-modulated transponder output signal through themedium of a pneumatic tire (e.g., 104).

RATIO VERSUS SIGNAL STRENGTH

An added advantage of using the ratio N_(T)/N_(P) for a pressureindicator accrues because it has been determined that the ratioed valueis less sensitive to variations in coupling between thereader/interrogator and the transponder than either of the N_(T) andN_(P) measurements taken alone. This is illustrated in FIG. 3D whichshows a graph 390 of measurement counts (on vertical axis 394) versuspower (on horizontal axis 392). For a passive transponder 200 such asdescribed in the preferred embodiment of this invention, the transponderpower is supplied by the RF signal from the reader/interrogator (e.g.,106). If the RF coupling strength weakens due to transmission orreception problems including excessive distance or interference, thenthe power in the transponder 200 circuitry may decrease. It has beendetermined that for power levels below a certain value PWR₁, therelaxation oscillator 312 outputs a lower than normal frequency signalFosc′ and thus reduces the temperature and pressure counts N_(T) andN_(P) below what they should be for a given temperature or pressure. Theeffect is illustrated by the downward curve on the plot 396 oftemperature count N_(T) and on the plot 398 of pressure count N_(P) asthe plots extend below the minimum power PWR₁. Fortuitously, thelow-power effect is proportionally the same for both counts, so that theratio N_(T)/N_(P) (plot 399) becomes relatively steady for all powerlevels down to a minimum power PWR₀ needed to operate the transponder200. Thus, by determining (during calibration) and storing calibrationdata for the ratioed value of N_(T)/N_(P) in the transponder, theability to determine a pressure-only reading which is relativelyinsensitive to coupling variations between the reader/interrogator andthe transponder is both simplified and made more reliable.

IMPROVEMENTS, GENERALLY

The present invention deals with a new “RFIQTM” transponder 400 (seeFIG. 4A) which implements improvements to the previous model, “3070C”transponder 200 described hereinabove. A number of improvements havebeen made, and new features incorporated, including, but not limited to:

Lower power consumption.

Increased oscillator stability vs. Power or Frequency.

Increased resolution of temperature and pressure counts.

Increased electrostatic discharge (ESD) protection to more than 2200V.

Increased, programmable modulation index (magnitude of PSK modulationapplied to the RF signal).

Reduced number of external connection pins for programming and testing.

Increased Digital and Analog Testability.

Increased to 192 bits of data stream.

Increased to 156 bits of programmable EEPROM.

Programmable with the antenna (coil) attached.

6 bits of parity −1 bit for each 4 bits of NT, NP data.

Programmably scale currents to independently optimize reading counts forpressure and temperature.

3V Battery powered mode for use in “active” implementations of thetransponder (“active tags”).

Power-On Reset.

Base-band data output on a test pin.

PROBLEMS TO BE SOLVED

In particular, there are general problems with the previous transponderdesign 200 described hereinabove. It is believed that improvements whichare the subject of the present invention provide solutions for a numberof these problems:

The relaxation oscillator of the previous design transponder 200 couldbe adjusted to optimize the pressure and temperature counts (NP and NT)for a desired pressure and temperature range, but the adjustment couldonly be accomplished during integrated circuit production. It isdesirable to provide a way to optimize the counts NP and NT afterintegrated circuit production.

When the previous design transponder 200 is used (as a passivetransponder), inaccurate results may be transmitted when the transponderoperates at too low a power input, such as during startup, or whendistant from the reader/interrogator. (For example, refer to FIG. 3D.)In addition, the previous model would start modulating (“transmitting”)as soon as it received an interrogation signal of any strength, andwould start the transmission at a random location in the data word. Ifthe initial signal was weak (from a distant reader/interrogator), thenthe voltage supplies could be insufficient to produce valid temperatureand pressure readings. And if modulation started before sufficient powerlevels were developed, then the power drain of signal modulation wouldaggravate the insufficient power problems. A further problem occurs ifthe relaxation oscillator does not start each measurement cycle in aconsistent, defined state.

The previous design transponder 200 has a fixed modulation index(magnitude of RF signal modulation) which is determined duringintegrated circuit production. It is desirable to provide a way tooptimize the transponder for different combinations of antenna (“coil”)210 and reader/interrogator 106, and for different operating conditions.Also, certain transponder applications utilize an external zener clampacross the antenna 210 for better stability, but the zener can cause theprevious design transponder 200 to be “read-limited”.

Although the previous design transponder 200 can have certain operatingcharacteristics adjusted (“trimmed”) during manufacture (e.g., currentscaling via size adjustments to transistors), these manufacturingchanges are permanent, and not easy to vary from transponder totransponder. Furthermore, if trim settings were to be implemented inprogrammable memory (e.g., EEPROM), then there would be other problemswith accessing those settings during power-up, and continuously ratherthan through row (242) and column (240) decoders.

Other improvements and problem solutions may become evident in thedescription to follow.

GENERAL DESCRIPTION

The improved RFIQ™ transponder 400 (compare 200, 102) is a custom CMOS,low voltage Integrated Circuit (“IC”, or “chip”) that can measuretemperature and pressure as a low-frequency “passive” (RFsignal-powered) transponder or as an “active” (battery-powered)measurement system. The IC provides programmable trimming thatdetermines: (a) if the part is active or passive, (b) to adjusttemperature and pressure resolution, and (c) modulation index and coilclamping strength (when in the passive mode).

In the passive mode, the IC develops power from a reader/interrogator RFsignal that is coupled to an external LC circuit across the transponderantenna inputs. The transponder uses the signal received to providepower and to generate an on-chip clock. The transponder sends back itsmemory contents to the reader by modulating the impedance of theantenna, which is known as “back scattered modulation”. The readerdemodulates the returned data to get the sensor identification (“ID”) aswell as sensor data and calibration constants needed to interpret thedata. The transponder sends a 196-bit serial phase-shift-keyed (PSK)data-stream as sixteen, 12-bit words. The first 12 bits are theprogrammable sync word, with each sync bit being 1.5 bits wide. Next,144 bits of EEPROM are transmitted that are of normal bit width. The 144bits contain the unique ID code of the transponder, the calibrationconstants for temperature and pressure data, and a CRC for errorchecking. The transponder then sends 36 bits of data: a 12-bittemperature count (NT), a 12-bit pressure count (NP), and a 12-bit wordconsisting of five unused bits as 1's, 1 bit for the state of the MTMSover-temperature sensor, and 6 bits of even-parity, with one bit ofparity for every four bits of NP and NT data.

In the active mode, the transponder must be controlled by externalhardware. The controlling hardware provides power to the IC and a clocksignal via either the CLK or VB pads. The IC shifts out its data on theDATA pad on every falling edge of the clock. The clocks must be given ata precise clock rate so that temperature and pressure are collectedduring fixed periods of time and therefore, the hardware can optimizethe clocking time per bit to get the highest resolution for temperatureand pressure. The IC can operate down to as low as 2.8V in active mode.

As either a passive or active tag (transponder), the IC can be testedafter assembly by applying power and talking to the chip via a 4-pininterface. This interface allows the user access to the chip's EEPROMand allows testing the chip functions. The EEPROM data can be read,cleared, or programmed and the oscillator tested by directly reading thefrequency. The chip sensor oscillator can also be tested and provides asecond method for reading temperature and pressure in the active mode.

OVERVIEW OF THE RFIQ™ TRANSPONDER CIRCUITRY

FIG. 4A is comparable to FIG. 2, and is a block diagram of a relevantportion of an improved RFIQ™ transponder 400 (compare 102, 200),illustrating the following signals, terminals and functional blocks(sections), and their interconnections with one another. This exemplarysystem is described as an embodiment which preferably measures pressureand temperature, but it is within the scope of the invention to includemeasurement of other parameters which employ suitable sensors.

The transponder 400 is preferably implemented on a single integratedcircuit (IC) chip shown within the dashed line 402 (compare 202), towhich are connected a number of external components. Other dashed linesin the figure indicate major functional “blocks” (“sections”) of thetransponder 400, and include a block 438 (compare 238) of addressablememory, and a sensor interface section 406 (compare 206).

The components external to the IC chip 402 include an antenna system 410(compare 210) comprising an antenna (coil) 412 and an optional capacitor414 connected across the coil 412 to form an L-C resonant tank circuit,an external precision resistor (Rext) 416 (compare 216), an externalpressure-sensing capacitor (C_(P)) 418 (compare 218), and an optionalexternal maximum temperature measurement switch (MTMS) 420 (compare220). Each external component has an appropriately labeled connectionpad as shown in FIG. 4A: VA and VB for the antenna system 410; Rext, Cpand MTMS for the high side of the precision resistor 416, thepressure-sensing capacitor 418, and the maximum temperature measurementswitch 420, respectively. The ground connection for the analog externalcomponents Rext, Cp, and MTMS should be made via the analog ground AGNDpad. The other ground pad (GND) is for the ground connection of externaldigital connections. The remaining connection pads are for use in activeor test modes of transponder 400 operation as described hereinbelow.

The antenna 412 may be in the form of a coil antenna, a loop antenna, adipole antenna, and the like. It is mainly used when the transponder 400is in the passive mode. Alternatively, when the transponder 400 is inthe active mode, the antenna system 410 may not be present, and thesignal output by the transponder 400 may be provided via directconnection to a DATA pad. In the main hereinafter, a transponder havinga coil antenna, and used in the passive mode is described.

The transponder IC 402 includes interface circuitry 422 (compare 222)for processing an RF signal, such as an un-modulated carrier signal offrequency Fi (e.g., 125 kHz) received by the antenna 412, and forrectifying the received RF signal so that it can be used to power thetransponder 400 operating in passive mode. The signal processingincludes passing on a suitable form of the incoming signal to be usedfor generating timing/clock pulses for the transponder 400, and alsoincludes applying modulation to the carrier signal for transmission bythe antenna system 410.

The rectified carrier signal is clamped to a maximum of approximately13.0 volts to prevent breakdown of the IC 402 substrate. The clamped,rectified signal has a voltage Vpp which can be read at a VPP pad, andranges from 0-13 volts. The Vpp voltage is then shunt-regulated to amaximum of 6.5 volts, and designated as supply voltage (or “inputvoltage”) Vxx which can be read at a VXX pad. The Vxx voltage levelfollows the Vpp voltage, and is about 6.2 volts for a typical receivedRF signal. Voltage Vxx is regulated to prevent voltages high enough topotentially damage CMOS circuitry in the IC 402.

The voltage Vxx is provided to a power-on reset (POR) circuit 482 (whichis new to this model transponder), and also to a regulation and bandgapreference circuit 423 (compare 222) for providing various voltagesupplies to the circuitry on the IC chip 402.

The power-on reset circuit 482 is provided to ensure that thetransponder 400 will not begin to record sensor readings or to transmitdata until enough power is being supplied by the received carrier signalto allow proper functioning of the transponder 400. The POR circuit 482evaluates the voltage Vxx level and outputs a reset signal which is notreleased until the Vxx voltage is deemed sufficient. If desired, such asfor test purposes, the reset signal can be imposed from outside the chip402 through an RES connection pad.

In the passive mode, as long as the supply voltage Vxx is sufficient (atleast 4 volts as determined by the POR circuit 482), the regulation andbandgap reference circuitry 423 will regulate voltage Vxx to provide aregulated analog supply voltage Vcc of approximately 3 volts, with anoperating range of 2.8 V (minimum voltage for a stable oscillator 452)to approximately 3.5 V. The digital supply voltage Vdd is regulated by asource follower connected to voltage Vcc and can supply currentindependently of voltage Vcc (compare with previous method as in FIG.3B, described hereinabove). Voltage Vdd is typically about one thresholdbelow Vcc, or approximately 2.5 volts, ranging from 1.2 V (minimumvoltage for stable logic and memory operation) to approximately 3.5 V,and supplies both the digital logic and the EEPROM (memory) array 436during reading. During transponder programming, voltages Vcc and Vdd arenot affected by applying power to pad VPP. The voltages Vcc and Vdd canbe read externally via their correspondingly-named connection pads (VCCand VDD). Also, in active (battery-powered) mode, the regulators can beoverridden by applying external power to the VCC and VDD pads. Finally,for circuits which need a stable reference voltage, the regulation andbandgap reference circuit 423 provides a temperature-independent bandgapvoltage Vbg. The reference voltage Vbg is also independent of the chipsupply voltage Vxx as long as the voltage Vxx is above a minimumoperating level. The regulation and bandgap reference circuit 423 willoutput a substantially temperature-independent voltage Vbg ofapproximately 1.20 volts over a transponder operating temperature rangeof, for example, −40 to 150 degrees C. The bandgap voltage Vbg is usedas a reference voltage by the Vxx, Vcc and Vdd regulators 423 as well asthe relaxation oscillator 452 and the power-on reset circuit 482.

The interface and rectification circuitry 422 also provides the receivedRF signal, preferably at the input frequency (Fi) it is received, to atiming and clock generator circuit 424 (compare 224, 226) whichgenerates clock signals in a known manner for controlling the timing ofother circuits on the IC chip 402. The generated clock signal is a 50%duty cycle square wave, preferably at frequency Fi, and is independentof any modulation applied to the antenna system 410 by the transponder400 for the transponder's PSK transmission. The timing and clockgenerator circuit 424 also divides down the system clock to developtiming for addressing of the data in the addressable memory 438, and forthe modulation. For example, the system clock frequency Fi is divided bytwo to determine the frequency of the PSK modulated return carriersignal. Other divisions of the frequency Fi are used for determining thebaud rate for data transmission. In active or test modes of operation,the timing and clock generator circuitry can be bypassed or used as abuffer for direct input of the clock signal via a CLK pad or the VBantenna pad.

From the timing and clock generator 424 the various clock signals arepassed to several control logic circuits: the column decoder 440(compare 240), the column to data converter 441, and the row decoder &N_(T), N_(P) control 442 (compare 242) which control access to datastored in the addressable memory 438. The clock signals are also used bythe row decoder & N_(T), N_(P) control 442 to control the timing of therelaxation oscillator 452 and the data capture circuitry 454, whichgenerate and store the temperature N_(T) and pressure N_(P) readings(counts) in the temperature register 432 and pressure register 434. Thusthe row decoder & N_(T), N_(P) control 442 functions as a “timinggenerator” secondary to the timing and clock generator 424.

The addressable memory block 438 includes an EEPROM array 436 (compare236) and several hardware registers 432, 434, and 435 (compare 232 and234). The EEPROM 436 is programmed with a variety of stored informationthat will be described in detail hereinbelow.

The last two rows of the EEPROM (e.g., rows 14 and 15) constitute atrimming bits section 436 b which is programmed to store trimminginformation. The trimming information: (a) controls the scaling of thecurrent (in the base-emitter voltage to current converter 450) suppliedto the relaxation oscillator 452 to optimize the pressure andtemperature count resolution, (b) sets the modulation index in themodulation circuit 446 to optimize signal transmission for a givenantenna system 410 and for a given transponder application, (c) sets themode of operation (active or passive), and (d) optimizes the impedanceof a clamp on the voltage Vpp across the rectification circuit 422.Utilizing the trimming lines 485, the trimming information stored in thetrimming bits 436 b can be directly read by the circuits it affects(e.g., 450, 446, 484, 422, 482) at any time during transponder operation(active or passive). Alternatively, as in certain test and programmingmodes, the trimming bits 436 b can be accessed for external reading andwriting (programming) along with the rest of the EEPROM 436 memory viathe DATA connection pad, as controlled by the test logic circuitry 484,and communicated by a transponder data line 444 through the column todata converter 441.

As described hereinabove with reference to the previous modeltransponder 200, the temperature and pressure registers (432 and 434respectively) are each a hardware register holding the count (e.g., 12bit) of a down-counter which is clocked by the sensor data signal offrequency Fosc′ coming from the sensor interface section 406.

The parity, MTMS 435 section of memory is also new to this model oftransponder 400. It is implemented as a register in hardware with, forexample, 12 bits of stored data.

Five bits are permanently set (value=1), then there is a bit whichsets/clears according to the open/closed (1/0) status of the MTMS switch(closes if exposed to excessive temperature), and finally six bitsregistering the parity of the pressure and temperature counts: threeparity bits for the pressure register 434 count N_(P), and then threeparity bits for the temperature register 432 count N_(T). The paritybits are continuously updated during sensor data capture, following thechanging counts in the N_(P) and N_(T) down-counters (the pressure 434and temperature 432 registers, respectively). Each parity bit representsthe parity of 4 sequential bits (a “nibble”) in the correspondingpressure 434 or temperature 432 register; with the most significantparity bit representing the most significant nibble of the count, themiddle parity bit for the middle nibble of the count, and the leastsignificant parity bit for the least significant nibble of the count.

The sensor interface portion 406 (compare 206) of the transponder chip402 consists of the base-emitter voltage to current converter 450(compare 250) with connection pad “Rext”; the relaxation oscillator 452(compare 252) with connection pad “Cp”; the data capture circuitry 454(compare 254); and the “MTMS” connection pad and line 459 connecting itto the MTMS bit in the parity, MTMS register 435 (compare 236 for theprevious location of the MTMS bit).

The base-emitter voltage to current converter 450 functions in a similarway to the converter 250 which is described in greater detailhereinabove with reference to sections 306 to 310 of FIG. 3. Forimproved performance, the circuitry of sections 306, 308 and 310 in thenew base-emitter voltage to current converter 450 utilizes cascodesinstead of single stage current mirrors (e.g., for transistors P1, P2,P4, P6), and the external resistor Rext 416 may have a differentpreferred resistance value, such as 500 kilohms. Cascodes are desirablebecause of their increased power supply rejection ration (PSRR). Also,in a feature to be described more fully hereinbelow with reference toFIG. 5, the final stage 510 (compare 310) of the base-emitter voltage tocurrent converter 450 is connected in a different way to section 308,and works in conjunction with programmed settings in the trimming bitsregister 436 b to provide a scaled proportional-to-temperature currentI(T) B on the line 451, 511 (compare current I(T)/N on line 251, 311) tothe relaxation oscillator circuitry 452 (compare 252). In contrast withthe previous design, the current scaling circuit 510 (compare 310) canscale the current I(T) by a variable scaling factor “B”, as determinedby programmed settings in the trimming bits register 436b.

The relaxation oscillator 452, under the timing control of the rowdecoder & N_(T), N_(P) control circuit 442, works in conjunction withthe external capacitive pressure sensor Cp 418 to generate a signal online 453 (compare 253) with frequency Fosc′ which is indicative ofeither the ambient temperature or the ambient pressure, depending on thetiming window determined by the row decoder & N_(T), N_(P) controlcircuit 442. The data capture circuitry 454, under the timing control ofthe row decoder & N_(T), N_(P) control circuit 442, directs the Fosc′signal to the appropriate hardware register (depending on the timingwindow): to the temperature register 432 via line 455, or to thepressure register 434 via line 457.

The current scaling circuit 510 (part of the base-emitter voltage tocurrent converter 450) and the relaxation oscillator 452 haveimprovements compared to the relaxation oscillator 252, and the MTMS 420status is directed to a different memory location (register 435) than inthe previous implementation (EEPROM 236). Otherwise, the sensorinterface section 406 functions essentially the same as thecorresponding section 206 in the previous model 3070C transponder 200.

As described hereinabove with reference to the previous modeltransponder 200, the term “ambient” refers to the parameter beingmeasured in the vicinity of the transponder 400, or more particularly inthe vicinity of the respective sensors associated with the transponder400. Also, references made herein to “pressure readings”, “pressurecounts”, “pressure response”, “pressure register” and the like generallyrefer to “pressure” as measured by this transponder technique whichactually produces a hybrid pressuretemperature reading. When this hybridreading has been processed to remove its temperature component, thereading is referred to as a “pressure-only” reading.

In conjunction with the column decoder 440 and the row decoder 442, thecolumn to data converter 441 controls the sequence in which signals(i.e., data) are output on a line 444 (compare 244) to a modulationcircuit 446 (compare 246) which, via the interface and rectificationcircuitry 422 (compare 222), communicates selected measured tireoperating characteristics in a data stream via the antenna system 410 toan external reader/interrogator (e.g., 106). The line 444 alsocommunicates the data stream to the test logic circuit 484 where it canbe directly accessed via the DATA connection pad.

The modulation circuit 446 converts the data stream from line 444 to arepresentative sequence of impedance changes (modulations) which areapplied to the antenna system 410 through the interface andrectification circuit 422. A new feature of the transponder 400 of thepresent invention is the ability to modify the modulation index(magnitude of modulation) to suit operating power levels and to select(via the trimming bits 436 b) a modulation index optimized for theindividual transponder 400, antenna system 410, and reader/interrogator106 in use.

In passive mode operation, an RF carrier signal from an external source(e.g., a reader/interrogator 106) is received by the antenna 412. ThisRF signal is rectified and used to power the RF transponder 400 as wellas providing the timing/clock signals. Modulating information applied bythe modulation circuit 446 is used to alter characteristics (e.g.,impedance, resonant frequency, etc.) of the LC tank circuit of theantenna system 410. These alterations are sensed as changes in load bythe external reader/interrogator 106 and are decoded, providingcommunication of data back from the RF transponder 400 to the externalreader/interrogator 106. Because the transponder 400 passive power isderived from the received RF signal, and because modulation of thatsignal drains off some of that power, the POR circuit 482 maintains areset signal during passive power-up, and will not clear the resetsignal (thereby allowing modulation), until the transponder power levelsare high enough to assure stable operation of the transponder 400 duringmodulation.

The test logic circuit 484 enables tests that can be performed at allphases of transponder production and use, including wafer sort, initialboard assembly level programming, programming at the pre-calibrationstage, calibration and trimming of the transponder, and post-calibrationprogramming to adjust the trimming bits 436 b for encapsulation-inducedoffset error.

More detailed explanations of significant portions of the RFIQ™transponder 400 of this invention are presented in the followingsections.

MEMORY ASSIGNMENTS AND THE DATA STREAM

The addressable memory block 438 is organized in a way which provides adata stream that is improved over that of the previous model 3070Ctransponder 200. The column decoder 440, the column to data converter441, and the row decoder & N_(T), N_(P) control circuit 442 worktogether to control the flow of data in and out of the addressablememory block 438. When operating in the active or passive modes (i.e.,not in a test or programming mode), the circuits 440, 441 and 442 accessthe memory locations one at a time in sequence from the first address tothe last (most significant bit to the least significant bit in each data“word”, with words sequenced from the lowest numbered word to thehighest), thereby producing a serial string of data for transmission. Itwill be seen from the following description that the rows to be includedin the data stream are selected according to the mode of operation(i.e., passive/active, or a variety of test and programming modes). Anadded function of the row decoder & N_(T), N_(P) control circuit 442 isto control the sensor interface circuitry 406 (via line 487) so that itis accumulating temperature-related counts in the temperature register432 during one portion of the data transmission (e.g., whiletransmitting words/rows 2 through 6), and pressure-related counts in thepressure register 434 during another portion of the data transmission(e.g., while transmitting words/rows 9 through 13), both accumulationsto be completed in time for the counts to be read back out of theregisters 432 and 434 when their part of the data stream is due (e.g.,words 14 and 15).

FIG. 4B (compare FIG. 3C) is a “map” of the addressable memory block438, showing its physical organization (by “rows”) and also showing theorganization of the active or passive mode data stream (by “words”). Inthe preferred embodiment of transponder 400 described herein, each wordor row is 12 bits (or columns) in length, and there are 16 words in thedata stream, comprising a total of 192 bits (12 times 16). The drawingof the memory block 438 in FIG. 4A is helpful in understanding thephysical organization. It should be noted that the rows 1 to 13 ofphysical memory (EEPROM 436) correspond to the words 1 to 13 of the datastream, but the rows 14 and 15 of physical memory (EEPROM 436) are notpart of the data stream. Instead, the temperature register 432 is readout as word 14, and the pressure register 434 is read out as word 15 ofthe data stream. The final word of the data stream, word 16, is read outof a hardware register of physical memory (the parity, MTMS register435).

The EEPROM 436 portion of the exemplary addressable memory block 438comprises 180 cells arranged in a 12 column by 15 row array. Each cellis made up of at least one n-channel select gate and one correspondingEEPROM transistor. The first 13 rows (sync, I. D., calibration, CRC 436a) are readable in the normal read modes whether the transponder 400 isprogrammed to be active or passive. The EEPROM memory locations in thesefirst 13 rows are selected in the usual way, with one n-channeltransistor row selection gate per EEPROM transistor. The EEPROM cells atrows 14 and 15 (trimming bits 436 b) hold the data for 12 bits oftrimming information, and are configured differently to facilitate theirspecial role in the transponder. Each trimming bits 436 b EEPROM cellhas two gates instead of one, adding a special READ_TRIM selection gatein series with the row selection gate so that the trimming bits 436 bcannot be added to the data stream unless the READ_TRIM selection gatehas been enabled as in certain test modes of transponder operation.Another feature of the special trimming bits 436 b register is that eachtrimming bits 436 b cell also has an added sensing line to communicatethe EEPROM bit setting (programmed trimming information) to appropriatesections of the transponder 400 circuitry as needed.

Thus the EEPROM 436 and associated logic are structured so that: Thetrimming bits 436 b data can be read externally by a “READ_TRIM” testmode which reads out the bits sequentially as words 14 and 15 in thedata stream in place of the temperature and pressure counts. In a “READ”test mode, as in normal operation, the trim bits are not seen in thedata stream, but are still accessed by sense amplifiers to communicateprogrammed trim settings to appropriate sections of the transponder 400.In a “WRITE” test mode, all the EEPROM 436 cells (including the trimbits 436 b) are addressed simultaneously and are written to. This ineffect writes ‘1’s to all the EEPROM cells, giving them high thresholds.In an “ERASE” test mode, individual cells can be erased (programmed to‘0’, given a low, negative, threshold): as the clock CLK signal indexesthrough the EEPROM cell array, a cell at the intersection of the columnaddressed and the row addressed is erased by raising the voltage on theVPP pad to the programming voltage and enabling the erasure by raisingthe DATA pad high.

Referring to FIG. 4B, the map illustrates an exemplary organizationwherein the first twelve data locations (bits 000 . . . 011 in row 1)are reserved for synchronization (“sync”) data. The next forty-eightdata locations (bits 012 . . . 059 in rows 2 through 5) are for generalinformation identifying the individual transponder 400. As mentionedhereinabove, each transponder unit is suitably calibrated prior to itsinstallation in a tire. The next twenty-four data locations (bits 060 .. . 083 in rows 6 through 7) hold temperature calibration data (e.g., adefining point and a slope). The next twenty-four data locations (bits084 . . . 107 in rows 8 through 9) hold pressure calibration data (e.g.,a defining point and a slope). The next thirty-six data locations (bits108 . . . 143 in rows 10 through 12) hold additional identifyinginformation concerning the IC chip 402. The next twelve data locations(in row 13) hold four bits of identifying information about the chipcalibration (bits 144. 147), and an eight bit value (bits 148 . . . 155)for a data validation algorithm such as CRC (Cyclic Redundancy Check).The next two words in the data stream (words 14 . . . 15, bits 156 . . .179) are read from the temperature register 432 and the pressureregister 434, respectively. The final word of the data stream (word 16,bits 180 . . . 191) is read from the parity, MTMS register 435, whichcontains five “open” bits (bits 180 . . . 184), then one bit (bit 185)containing the logic level (state) of the MTMS switch 420, and finallysix bits (bits 186 . . . 191) containing 3 bits each for the parity ofthe pressure count followed by the parity of the temperature count. Thefive open bits are unused, and are fixed as logical “1” values.

The column decoder 440, column to data converter 441, and row decoder &N_(T), N_(P) control 442 circuits coordinate addressing and access tothe addressable memory block 438.

The column decoder 440 consists of a four-bit synchronous counter thataddresses a 1-of-12 decoder. The outputs from the column decoder 440address the 12 columns of the memory array during programming andreading. To address a column, the address n-channel device routes a lowpower current-source to charge the addressed column. If a bit is writtento, the column will be pulled high, but if erased, the column will bepulled low. The output of the current-source is buffered and drives atransponder data line 444 that goes through the column to data converter441 to the modulation 446 and test logic 484 circuits. The columns aresequentially addressed from column 1 to column 12 at the rate of atiming logic clock signal provided either by the timing and clockgenerator 424, or externally via the CLK pad. After addressing column12, the column decoder 440 triggers the row decoder & N_(T), N_(P)control 442 to clock it to the next row and also cycles itself back tocolumn 1. Whenever a reset signal (turned on, then off) is received fromthe power-on reset circuit 482 (or via the RES pad), the column decoder440 and the row decoder & N_(T), N_(P) control 442 will reset theiraddressing sequence to start at column 1 of row 1, i.e., the first bitor cell in the addressable memory block 438. The column decoder 440serially addresses the EEPROM 436 array in any of the reading modes ofoperation or during erasing. In the WRITE test mode it has no effect,since the entire EEPROM memory is simultaneously addressed for the WRITEoperation.

The row decoder & N_(T), N_(P) control 442 is a 4-bit synchronouscounter that addresses a 1-of-16 decoder. The decoder addresses 13 rowsof EEPROM data memory 436 a from rows 1 though 13. In the WRITE, ERASE,and READ_TRIM test modes, it also addresses rows 14 and 15 of the EEPROMmemory 436 b for the trimming bits. During normal reading (active orpassive mode), the decoder addresses the temperature 432 and pressure434 hardware registers at row addresses 14 and 15, respectively.Regardless of operating mode, the row-16 address is directed to aspecial data row: the parity, MTMS 435 hardware register. In the READand ERASE test modes each row is addressed sequentially on the fallingedge of the last bit of the column decoder 440. In the passive or activemode, a reset signal sets the row decoder & N_(T), N_(P) control 442 torow 1, which is the sync word. In the WRITE mode, all the rows aresimultaneously addressed, which pulls all select lines and control gatesto voltage Vpp. As mentioned hereinabove, an added function of the rowdecoder & N_(T), N_(P) control circuit 442 is to control the sensorinterface circuitry 406 so that it is accumulating temperature-relatedcounts in the temperature register 432 and pressure-related counts inthe pressure register 434 during their designated time periods (datacollection windows).

PROGRAMMABLE CURRENT SCALING

As described hereinabove for the previous model 3070C transponder 200,the scaled current I(T)/N which is input to the relaxation oscillator312, 252 on line 311, 251 must be scaled by a fixed amount N determinedby a ratio of physical areas for the current mirroring transistors suchas P5 and P6 in the current scaling circuit 310 of FIG. 3. The physicalareas were set during fabrication.

A portion of FIG. 5 illustrates an exemplary improved (programmable)current scaling circuit 510 which is part of the base emitter voltage tocurrent converter circuit 450 of the RFIQ™ transponder 400 of thisinvention. The current scaling circuit 510 (compare 310) develops ascaled current I(T)B on line 511 (compare 311) wherein the scalingfactor “B” can be programmed into the transponder 400 via certain of thetrimming bits 436 b at any time after transponder 400 fabrication. Thecurrent scaling is now separately programmable for the temperaturemeasurement (e.g., temperature scaling factor B_(T)=1 or 1.5), and thepressure measurement (e.g., pressure scaling factor B_(P)=1 to 8.5 in0.5 steps), making the current supplied to the relaxation oscillator 512(compare 312) programmably increasable up to 1.5 or 8.5 times theproportional-to-ambient-temperature (PTAT) current I(T). By programmingthe current mirror in the current scaling circuit 510, the counts perdegree or counts per PSI can be maximized programmably, therebyincreasing the resolution and stability for temperature and pressurecounts (NT and NP). As detailed hereinabove, it is particularlydesirable to optimize the pressure count, therefore a large range ofpressure scaling factors B_(P) are made available for programming(trimming). The lesser range of temperature scaling factors B_(T) isgenerally adequate to compensate for shifts in processing at the dielevel, as well as changes in the external precision resistor (REXT) 416.It is within the scope of this invention to provide, in a similarfashion, other suitable programmable scaling factor amounts for both thetemperature B_(T) and pressure B_(P) scaling factors.

Referring to FIG. 5, the programmable current scaling circuit 510 isbounded by a dashed line. The transistor P6 is connected in acurrent-mirroring configuration with the transistors P1, P2 and P4 inthe temperature sensing and current mirroring circuitry (comparable to306 and 308 in FIG. 3, but only transistors P4 and N5 are shown in FIG.5) such that these current mirror gates are connected via line 505(compare 305) which supplies a reference voltage Pbias to the gate oftransistor P6. Because of the current mirror connection, the currentthrough transistors P4 and P6 will mirror the PTAT current I(T) throughthe external resistor Rext 416, 216. The voltage supply for theprogrammable current scaling circuit 510 is voltage Vcc provided on line503 (compare 303). As explained hereinabove, the voltage Vcc isregulated and substantially independent of temperature. The use of theanalog supply voltage Vcc is an improvement over the previous (circuit310) use of the digital supply voltage Vdd, because the voltage Vcc is“cleaner,” not having any digital switching noise imposed on it. For thesake of simplicity, the transistors P4, P6, P6.05, P6.1, P6.2, and P6.4are illustrated as single transistors. It is within the scope of thisinvention to implement these transistors as cascodes, as mentionedhereinabove for the transistors such as P1, P2 and P4 of the circuits306 and 308 of FIG. 3. It should be understood that such cascodetransistors used in a current-mirror arrangement would have separatebias lines (e.g., Pbias would be split into Pbias′ and Pbias″)connecting the gates of the transistors in each stage of the cascodecurrent mirror.

As in the current scaling circuit 310, the transistors P4 and P6 areconnected in a current-mirroring configuration. However, instead of thecircuit 310 transistors P5 and P6 having a fixed ratio of physical areasto scale the mirrored current, in the programmable circuit 510 of thisinvention the transistor P6 has a programmable ratio of physical areascompared to the transistor P4 (and transistors P1, P2 of circuit 306 and308) with which P6 is mirror-connected. The physical area of transistorP6 is changed by additional scaling transistors (e.g., P6.05, P6.1,P6.2, and P6.4) which are added in parallel to transistor P6, andswitched into use by switches under the control of certain of thetrimming bits 436 b and also control signals from the row decoder &N_(T), N_(P) control circuit 442. It should be understood that each ofthe transistors P4, P6, P6.05, P6. 1, P6.2, and P6.4 illustrated in FIG.5 has a suitable physical area which could be determined by the size ofan individual transistor, or could be the result of adding together thephysical areas of multiple transistors combined in parallel andcollectively labeled as “a single transistor”. For example, transistorP5 which has a physical area of 1 (in arbitrary “units”), could befabricated as one transistor with a physical area of 1.0 unit, or as twotransistors in parallel, each having a physical area of 0.5 units. Inthe embodiment illustrated in FIG. 5, the transistors P4, P6, P6.05,P6.1, P6.2, and P6.4 each have their relative physical areas labeled as“A=”n, where n is the physical area in arbitrary units. The transistorsP4, P6, and P6. 1 each have a physical area of 1.0 unit, and transistorsP6.05, P6.2, and P6.4 have physical areas of 0.5, 2.0, and 4.0 units,respectively. It should be understood that since the mirroredtransistors P1, P2 and P4 of FIG. 3 also have the same relative size asP6, then they also have a physical area of 1.0 unit. By switching invarious combinations of these added transistors P6.05, P6.1, P6.2, andP6.4 (“P6.n”), combined physical areas from 0.0 to 7.5 units can beadded to the area (1.0 units) of transistor P6 in increments of 0.5units. As explained hereinabove, the ratio of physical areas of themirrored transistors (e.g., the ratio (P6 +P6. n)/P4) will cause themirrored current I(T)B exiting the programmable current scaling circuit510 on line 511 to be scaled up from PTAT current I(T) by a factor Bwhich equals the ratio of total physical areas.

Each of the added transistors P6.n has one or more controllingsemiconductor switches in series (scaling trim switches S5, S6, S7, S8,and S9; temperature scaling switch ST1; and pressure scaling switchSP1), illustrated herein as a box with the control input indicated by aline on one side. Two kinds of semiconductor switches are illustrated:inverting and standard (non-inverting). The switches ST1 and SP1 arestandard semiconductor switches which could be implemented, for example,as N-channel transistors with the control signal going to its gate. Sucha switch will conduct (switch “on” or “closed”) when the control signalis high (voltage above ground such as logical “1”, “true”, or “set”),and will not conduct when the control signal is low (ground, logical“0”, “false”, or “cleared”). The switches S5, S6, S7, S8, and S9 areinverting semiconductor switches as indicated by the small circle at thecontrol input. These switches could be implemented, for example, asP-channel transistors with the control signal going to its gate. Such aswitch will conduct (switch “on” or “closed”) when the control signal islow, and will not conduct when the control signal is high. The invertingswitches S5, S6, S7, S8, and S9 are programmed “on” (closed) if thecorresponding trim bit 5 to 9 is programmed as “cleared”. It may benoted that in the embodiment of transponder 400 herein described, thetrim bits 5-9 are the only “low-true” logic bits in the trimming bits436 b register. An advantage of using such P-channel transistor switchesis that they will function sooner than N-channel transistor switches asvoltage levels increase during transponder power-up. The trim bits arebit addresses in the EEPROM trimming bits register 436 b, communicatedvia lines 485. In this example, trim bit 5 controls temperature countfine trimming, trim bit 6 controls pressure count fine trimming, andtrim bits 7, 8, 9 control pressure count gross trimming. The setting fora scaling trim bit “n” is illustrated as a scaling trim bit controlsignal labeled “TRIMBIT_”n (i.e., scaling trim bit signals TRIMBIT_5 . .. TRIMBIT_9 for scaling trim bits 5 through 9). The non-invertingscaling switches ST1 and SP1 are controlled on/off by the signalsCAPTURE_N_(T) and CAPTURE_NP, respectively, coming on lines 487 from therow decoder & N_(T), N_(P) control circuit 442. The scaling switches ST1and SP1 enable different scaled currents I(T)B to be used by therelaxation oscillator 452, 512 during the temperature measurement orpressure measurement periods (time windows WT and Wp, respectively).Thus the current scaling factor B will have a value B_(T) (1 or 1.5)during the temperature measurement window W_(T) (signal CAPTURE_NT ison), and will have a value B_(P) (1 to 8.5 in 0.5 increments) during thepressure measurement window W_(P) (signal CAPTURE_NP is on). AlthoughFIG. 5 illustrates a serial arrangement of switches (e.g., S5 and ST1),it is within the scope of this invention to avoid accumulated switchinglosses in the I(T)B signal by using digital logic to combine controlsignals to control a single switch.

For example, the CAPTURE_NT signal could be combined with an invertedTRIMBIT_5 signal by an AND gate, whose output would control switch ST1,thereby eliminating the switch S5.

RELAXATION OSCILLATOR

As described hereinabove for the previous model 3070C of transponder200, the relaxation oscillator circuit 312, 252 produces a signal offrequency Fosc′ which is determined by the alternate charging of twocapacitances: C_(FX1) and C_(FX2) for temperature readings, or C_(FX1)and (C_(FX2) Plus C_(P)) for pressure readings. The charging rate of thecapacitances is determined by the magnitude of the capacitances and themagnitude of the scaled current I(T)/N used to charge them. Thedischarge rate of the capacitances is not a factor because eachdischarges while the other is charging, and they discharge more rapidlythan they charge due to low resistance discharge paths. The set-resetpart of the relaxation oscillator 312 is triggered to flip-flop onlywhen one of the capacitances charges to a voltage level just above thebandgap voltage Vbg.

A portion of FIG. 5 illustrates an exemplary improved relaxationoscillator circuit 512 (compare 312) which substantially constitutes therelaxation oscillator 452 of the RFIQ™ transponder 400 of thisinvention.

The circuit 512 is still a relaxation oscillator, and is driven by ascaled current (the measurement current) on line 511 (compare 311) froma now-programmable current scaling circuit 510 (compare 310) asdescribed hereinabove. The set-reset portion 514 (compare 314 a and 314b) of the relaxation oscillator 512, with phase paths φ1 and φ2 (phaseone path 514 a and phase two path 514 b, respectively), functions in thesame general way but has been slightly modified in known ways toaccommodate the changes to the inputs of the comparators 516 (516 a, 516b, compare 316 a, 316 b). The major circuit changes are to the front-endswitching arrangement which has been divided into two measurementswitching circuits 515 a and 515 b for temperature and pressuremeasurements, respectively. The capacitor C_(FX2) has been eliminated,so that now the capacitor C_(FX) (compare C_(FX1)) is used exclusivelyduring the temperature measurement period, and the pressure sensingcapacitor C_(P) (418, compare 218) is used exclusively during thepressure measurement period. During a measurement period, the relaxationoscillator output signal frequency Fosc′ is now determined by thealternating charging and discharging of a capacitor C_(FX) or C_(P),whichever is selected by the row decoder & N_(T), N_(P) control circuit442. With the comparators 516 a and 516 b having reference voltages of,for example, Vbg and Vbg/2, respectively (the bandgap voltage and halfthe bandgap voltage), the selected capacitor C_(FX) or C_(P) will usethe (scaled) measurement current I(T)B to be charged up to just abovevoltage level Vbg, which will trigger the set-reset circuit 514 toflip-flop from its first state (e.g., PHASE1=true) to a second state(e.g., PHASE2=true), which switches off the charging current I(T)B andinstead diverts the scaled current I(T)B through a current mirror whichcauses the selected capacitor C_(FX) or C_(P) to be discharged by amirrored current equivalent to I(T)B from voltage level Vbg down to justbelow a voltage level Vbg/2, which will trigger the set-reset circuit514 to flip-flop again to the first state and begin charging theselected capacitor C_(FX) or C_(P) again. It can be seen that, after theinitial charging from ground (zero volts) to voltage Vbg/2, thedescribed operation will result in a uniform 50% duty cycle waveformwith a frequency which is determined by the magnitudes of the scaledcurrent I(T)B and the selected capacitor C_(FX) (for temperaturemeasurement) or C_(P) (for pressure measurement). Another new feature ofthe relaxation oscillator 512, 452 of this invention is the provision ofa small bias current Ibias which is switched in and mirrored to slowlydrain-to-ground either of the capacitors C_(FX) or C_(P) when they arenot selected for measurement use, and this grounding of the capacitorswhen not in use is also used to set the set-reset circuit 514 in adefined state before each time it is used for a measurement.

The exemplary relaxation oscillator circuit 512 has two analog inputs:the (programmably) scaled PTAT current I(T)B supplied on lines 511, andthe bias current Ibias supplied on lines 513. The bias current Ibias isa small fraction of the PTAT current I(T) (e.g., I(T)/10) derived in aknown way, such as the current scaling procedures described hereinabove.

The exemplary relaxation oscillator circuit 512 has two digital (logicor control) inputs supplied on lines 487 from the row decoder & N_(T),N_(P) control circuit 442: the CAPTURE_NT signal and the CAPTURE_NPsignal which determine whether temperature or pressure, respectively, isbeing measured. Control signal inputs are indicated in several places bytheir corresponding number. For example, the CFX_TO_OSC signal isindicated by a number one in a circle where it is output from the LOGICcircuit 531 and also where it is input to switch ST2. The outputs of thephase paths φ1 and φ2 are used as control signals PHASE1 and PHASE2,respectively, and are also inverted and used as control signals: thephase path φ1 signal is inverted by the inverter 520 a to produce thesignal {overscore (PHASE1)}, and the phase path φ2 signal is inverted bythe inverter 520 b to produce the signal {overscore (PHASE2)}. The lineabove the name herein indicates that the logic is inverted, or “NOT”, sothat, for example, {overscore (PHASE1)} means “not phase 1” (i.e., whenthe PHASE1 signal is high, or logic level 1, the {overscore (PHASE1)}signal will be low, or logic level 0). The PHASE1, PHASE2, {overscore(PHASE1)} and {overscore (PHASE2)} signals are combined with theCAPTURE_NT and the CAPTURE_NP signals as inputs to a logic circuit 531which outputs six control signals-two capacitor selection signals:CFX_TO_OSC (#1), and CP_TO_OSC (#2); and four measurement controlsignals: PHASE1_NT (#3), PHASE2_NT (#4), PHASE1_NP (#5), and PHASE2_NP(#6).

The details of the digital logic included in the logic circuit 531 areshown as the exemplary logic circuits 531 a . . . 531 f (for the signals(#1) . . . (#6), respectively) illustrated in FIG. 5A. Each of the logiccircuits 531a . . . 531f utilizes a NAND gate and one inverter toproduce its control signal to be output from the logic circuit 531. Itis within the scope of this invention to include other forms of logiccircuitry 531 which generate signals (#1) to (#6) which behave asdescribed herein. It can be seen that the output signals (#1) and (#2)generally signify that it is time to measure one parameter (N_(T) orN_(P)) and not the other (N_(P) or N_(T)). Note that these signals willalso be true (logical 1) whenever it is not time to measure the oppositeparameter, regardless of the setting of the “capture . . . ” signal.Thus, the signal CFX_TO_OSC (#1) is true whenever the signal CAPTURE_NPis false; and the signal CP_TO_OSC (#2) is true whenever the signalCAPTURE_NT is false. Since the (#1) and (#2) signals control theswitches ST2 and SP2, respectively, it can be seen that this logiccauses the switch ST2/SP2 to connect the capacitor CFX/CP to theset-reset circuit 514 whenever the opposite capacitor CP/C_(FX) is notconnected to the set-reset circuit 514. This helps keep the set-resetcircuit 514 in a defined state at all times.

Each of the output signals (#3) to (#6) generally indicate the state ofa signal which primarily determines, when true, which phase theoscillator is in (phase 1 or phase 2) and also which capacitor is in usefor a measurement, if any (C_(FX) for temperature, or C_(P) forpressure). As can be seen in the illustrated exemplary NAND gate logicin FIG. 5A, each signal (#3) to (#6) will also be true whenever theopposite measurement is selected, or even if no measurement is selected,regardless of phase. For example, the NAND gate in logic circuit 531 coutputs the signal PHASE1_NT (#3). If temperature is being measured,then PHASE1_NT will be high during phase 1 (charging C_(FX)) and lowduring phase 2 (discharging C_(FX)). However, when temperature is NOTbeing measured (i.e., CAPTURE_NT is low, not true), then PHASE1_NT willbe high regardless of the phase of the oscillator, and even if neithertemperature nor pressure are being measured. Similar logic applies toeach of the four control signals (#3) to (#6), designated by theencircled numbers #3, #4, #5, and #6, and illustrated in FIG. 5A aslogic circuit 531 c to 531 f.

The {overscore (PHASE)}1 signal, on line 521 (same as 453, compare 321and 253) is also the 30 (inverted) relaxation oscillator output signalFosc′ which is used by the transponder 400 to accumulate the temperatureand pressure counts in the corresponding registers 432 and 434,respectively. It can be seen that the relaxation oscillator 512 outputFosc′ is buffered and level-shifted to the digital logic signal level.The relaxation oscillator 512 can be tested or read directly by placingthe transponder 400 in either of the (active mode) test states,“READ_TEMP” and “READ_PRESS”. These test states enable the relaxationoscillator 512 for temperature or pressure, respectively, and direct therelaxation oscillator 512 signal Fosc′ to the DATA pad.

The front end of the relaxation oscillator 512 consists of twoessentially duplicate measurement switching circuits 515 a and 515 b,which utilize the temperature or pressure measuring capacitor, C_(FX) orC_(P) respectively, and a series switch ST2 or SP2, respectively, toconnect the appropriate front end measurement switching circuit 515 tothe set-reset circuit 514. As described hereinabove for switches ST1 andSP1, the switches ST2 and SP2 are standard (non-inverting) semiconductorswitches, such as could be implemented using N-channel transistors, orin CMOS. The capacitors on chip 402 are poly-to-poly capacitors, whichhave very low temperature coefficients and have areas (capacitance)which give high resolution, but with low sensitivity to processingfactors such as etching and mask alignment errors. The temperaturemeasuring capacitor C_(FX) is a fixed poly-to-poly capacitor of, forexample, 6 pf (+/−10%). The pressure measuring capacitor C_(P) 418(compare 218) is off-chip and preferably has similarly rugged and stablecharacteristics as described hereinabove (e.g., a touch mode capacitivepressure sensor with a linear capacitance versus absolute pressureresponse, varying between 4-40 pf).

The temperature measurement switching circuit 515 a consists, forexample, of P and N-channel transistors P9, P10, P13, N9, N10, N13, andN14; an inverter 517 a, a semiconductor switch ST2 (such as an N-channelCMOS transistor), and the temperature measuring capacitor C_(FX). Thepressure measurement switching circuit 515 b consists, for example, oftransistors P11, P12, P14, N11, N12, N15, and N16; an inverter 517 b; asemiconductor switch SP2, and the pressure measuring capacitor C_(P)(418, external to transponder chip 402 and connected via connecting padCp). Various control inputs, described hereinabove, are connected tothese measurement switching circuits 515 as indicated by the encirclednumbers 1 to 6, and the currents Ibias and I(T)B are also input whereshown.

The functioning of the temperature measurement switching circuit 515 awill now be described for illustration of the techniques involved. Thepressure measurement switching circuit 515 b functions in a suitableparallel fashion.

When the transponder 400 places the relaxation oscillator 512 in atemperature measuring mode, the row decoder & N_(T), N_(P) controlcircuit 442 will set the CAPTURE_NT signal high (and the CAPTURE_NPsignal low) for the time period t_(T) it chooses as the temperaturemeasurement window W_(T), thereby closing the switch ST2 (and holdingswitch SP2 open) via the logic circuits 531 a and 531 b. The set-resetcircuit 514 will start operation in phase 1 (i.e., the output of φ1, 514a is high; and the output of φ2, phase 2, 514 b is low) because wheneverthe relaxation oscillator is not being used for a measurement, themeasurement switching circuits 515 are always forced to a low voltage(as will be explained hereinbelow) and this low voltage is presented tothe comparators 516 a and 516 b due to the logic for signals #1 and #2as explained hereinabove. With the signals PHASE1=true and PHASE2=false,the logic circuit 531 will output the following signals, identified bytheir encircled numbers: PHASE1_NT (#3), PHASE1_NP (#5), and PHASE2_NP(#6)=true; PHASE2_NT (#4)=false. These signals cause transistors P13 tobe off and P9 to be on, thereby directing the scaled current I(T)Bthrough P9 to charge the capacitor C_(FX). Since switch ST2 is closed bysignal (#1), the set-reset circuit 514 can sense the rising voltage onthe capacitor C_(FX). At the same time, transistor N14 is held on,grounding the gate of transistor N9 and holding it off so that there isno bleed-off path for the capacitor C_(FX) being charged. Sincetransistors P10 and N13 are also held off, there is no other input tothe gate of transistor N9, and the state of transistor N10 is of noconsequence.

When the set-reset circuit 514 flip-flops to phase 2 (PHASE1=false,PHASE2=true), the signals PHASE1_NT (#3) will change to false, andPHASE2_NT (#4) will change to true (but PHASE1_NP (#5), and PHASE2_NP(#6) will remain “true”). These signals cause transistor P13 to be onand P9 to be off, thereby directing the scaled current I(T)B totransistor N10. Since transistor N13 is switched on, and transistor N14is switched off, the transistor pair N10/N9 now forms a current mirror,so that the current passing through transistor N10 to ground is mirroredby an equivalent current draining the capacitor C_(FX) throughtransistor N9 to ground. An unintended side effect of connecting thecurrent Ibias through the now-turned-on bias control transistor P10 isthat the current through transistors N10/N9 will be the sum of thescaled current I(T)B and the bias current Ibias, thereby slightlyincreasing the discharge rate of the capacitor C_(FX) compared to thecharging rate. Since the bias current Ibias is relatively small, thiswill change the relaxation oscillator 512 duty cycle to be slightly offof 50%—50%, but will not affect the counting process. (Compare theprevious model transponder 200, described hereinabove, which operatedwith an intentionally asymmetric duty cycle.) Since switch ST2 is stillclosed by signal (#1), the set-reset circuit 514 can sense thedecreasing voltage on the capacitor C_(FX).

When the temperature measurement window W_(T) is closed (transponder 400is no longer in a temperature measuring mode), the row decoder & N_(T),N_(P) control circuit 442 will set the CAPTURE_NT signal low, causingthe data capture circuit 454 to turn off data acquisition in thetemperature register 432. The switch ST2 stays closed at this point(signal CFX_TO_OSC=1) because the CAPTURE_NP signal is still low. Theswitch ST2 will only open at another time if the CAPTURE_NP signal isset high (and the CAPTURE_NT signal is low), i.e., pressure is beingmeasured and not temperature. With the CAPTURE_NT signal low, the logiccircuit 531 will cause both the signals PHASE1_NT (#3) and PHASE2_NT(#4) to be true. These signals cause both transistors P13 and P9 to beoff, thereby cutting off the flow of the scaled current I(T)B throughthe temperature measurement switching circuit 515 a. However, transistorN13 is switched on, and transistor N14 is switched off, so that thetransistor pair N10/N9 still forms a current mirror, and the biascontrol transistor P10 is on so that the bias current Ibias passingthrough transistor N10 to ground is mirrored by an equivalent currentdraining the capacitor C_(FX) through transistor N9 to ground. Since the(small) bias current Ibias is the only current allowed to flow throughthe mirror circuit N10/N9, the capacitor will be gradually drained toground (0 volts) and held there until the next temperature measuringwindow W_(T) is opened by making the CAPTURE_NT control signal high(true). It should also be noted that since the switch ST2 is stillclosed, the grounded inputs to the comparators 516 a and 516 b willforce the set-reset circuit 514 to stop oscillation in the defined stateof PHASE1=high, PHASE2=low. It is a feature of this invention to providethe herein described means of utilizing a small bias current lbias toreset the measurement capacitors C_(FX) and C_(P) to defined states(zero-voltage) before each usage of the measurement capacitors C_(FX)and C_(P) to accumulate temperature counts N_(T) or pressure countsN_(P), respectively. It is a further feature of this invention to usethe herein described means to additionally place the set-reset circuit514 in a defined state before each use of the set-reset circuit 514.

It may be noted that the exemplary description hereinabove utilizes thecontrol signals CAPTURE_NT and CAPTURE_NP for different purposes indifferent portions of the transponder 400 circuitry, e.g., in thebase-emitter voltage to current converter 450 (current scaling circuit510), in the relaxation oscillator 452 (512), and in the data capturecircuit 454 (substantially the same as the circuit 254 illustrated inFIG. 3A). It is within the scope of this invention to modify the timingof the control signals to each of these circuits 450, 452, and 454 sothat the events occur in a suitable sequence. For example, the circuit454 may have a delayed turn-on (e.g., delay 1 bit-width time of 256μsec) to allow the relaxation oscillator 512 time to reach stableoperation, oscillating between the voltages Vbg/2 and Vbg on themeasurement capacitors C_(FX) and C_(P).

As noted hereinabove for the typical passive mode operation of thetransponder 400, the row decoder & N_(T), N_(P) control circuit 442provides the control signals to the sensor interface circuitry 406 (vialine 487) so that it is accumulating temperature-related counts in thetemperature register 432 during one portion of the data transmission(e.g., while transmitting words/rows 2 through 6), and pressure-relatedcounts in the pressure register 434 during another portion of the datatransmission (e.g., while transmitting words/rows 9 through 13). Toimplement a delayed turn-on for the data capture circuit 454, the rowdecoder & N_(T), N_(P) control 442 could, for example, send theCAPTURE_NT signal to the programmable current scaling circuit 510 andalso to the relaxation oscillator circuit 512 upon clocking to bit 1 ofrow 2, and then send a different “Capture Temp” signal to the datacapture circuit 454 upon clocking to bit 2 of row 2. In like fashion,the CAPTURE_NT signal and the “Capture Temp” signal could be sent atdifferent times, or even together (e.g., upon clocking to bit 1 of row7).

ADJUSTING TEMPERATURE RESPONSE

It can be seen from the description hereinabove that the transponder200, 400 utilizes an inventive temperature sensing circuit 306 (see FIG.3) for the measurement of ambient temperature, which measurement is usedby the transponder 200, 400 to produce a temperature measurement outputsignal in the form of temperature counts N_(T). In the embodimentdescribed hereinabove the temperature sensing circuit 306, utilizing alateral bipolar transistor Q1 as a temperature sensor and an externalresistance Rext as a voltage-to-current converter, is optimized for usein transponders monitoring pneumatic tire conditions by selecting forthe resistance Rext 216, 416 a precision resistor with a value of about455 kilohms. Testing of a transponder 400 having a precision resistorRext 216, 416 with a fixed value of 453 kilohms produced a reasonablylinear temperature count N_(T) versus temperature plot with an overallslope of approximately −7 counts per degree C (−3.89 counts/° F.) whichallowed measurement of temperature over the full desired temperaturerange of −20° C. to 100° C. (−4° F. to 212° F.) for truck tires. Theresolution of the measurement is essentially the inverse of the slope,i.e., degrees per count, since the transponder output of temperaturecounts N_(T) is an integer which cannot be resolved to less than onecount. For truck tire temperature measurement {fraction (1/7)}° C. (oneseventh of a degree Celsius) is considered adequate resolution. Over the120° C. range, this slope/resolution only utilizes 840 counts which iswell within the 4095 count available count range. A steeper slope, thusbetter resolution, could be obtained by (as explained hereinbelow)reducing the resistance value of resistor Rext 216, 416, but that wouldraise the current to an unacceptably high level for the transponders 200and 400 which are designed to be low-power, primarily passive devices.In fact, the minimum resistance recommended for resistor Rext 216, 416in the newer transponder 400 is 200 kilohms. This presented a problem insituations where the transponder 400, or even the temperature sensingcircuit 306 by itself, is desired to be used for temperaturemeasurements requiring better resolution, e.g., animal body temperaturemonitoring, where the range is smaller (95° F. to 105° F., or 35° C. to41° C.), but much higher resolution is desired.

To solve the resolution problem without producing unacceptable currentlevels in the temperature sensing circuit 306 or the transponder 200,400, the resistance of resistor Rext 216, 416 was provided by athermistor 716 either replacing or supplementing the fixed resistance ofa precision resistor. FIG. 7 illustrates a response-adjustingtemperature measuring device circuit 700 including the temperaturesensing circuit 306 of FIG. 3 with power supply Vcc 303, Pbias line 305and startup circuit connection 307, but the external resistance Rext isherein provided by a thermistor 716 instead of the fixed resistor 216 ofFIG. 3. Since the temperature sensing circuit 306 is an integral part ofthe base-emitter voltage to current converter 250, 450 in thetransponder 200, 400, it should be understood that the externalresistance Rext labeled 216 and 416, respectively, in the transponder200, 400 schematics of FIGS. 2 and 4A would likewise be replaced orsupplemented by the thermistor 716 if the inventive circuit 700 isemployed in the transponder 200, 400. Although not illustrated, itshould be apparent that alternative embodiments of the circuit 700 couldinclude fixed resistors added either in parallel or in series with thethermistor 716 in order to supplement the resistance value of theexternal resistance Rext. Therefore the previous mentionedsupplementation is considered within the scope of this invention.

The rational behind the inventive utilization of the thermistor 716 liesin the need for a change in the slope of the temperature response. Asexplained hereinabove, the temperature-indicative output of thetemperature sensing circuit 306 is a PTAT(proportional-to-ambient-temperature) current I(T) mirrored by way ofthe Pbias line 305 (also 505). In transponder 200, 400 applications, themirrored PTAT current is scaled by a temperature scaling circuit 310,510 and then utilized to drive a relaxation oscillator 252, 452, 312,512 which outputs a signal Fosc′ which has a frequency proportional tothe (scaled) PTAT current, and which is then used to produce atemperature measurement count N_(T), also proportional to the PTATcurrent. The mirrored PTAT current I(T) is equal to the current throughthe external resistance Rext 216, 416, 716, and is determined by theemitter voltage of the transistor Q1 divided by the value of theexternal resistance Rext. The emitter voltage of the CMOS lateralbipolar transistor Q1 (due to its base-emitter voltage difference)inherently varies with temperature by a constant, predictable amount(temperature coefficient) of −2.2 mv/° C. By Ohm's law, the current I(T)equals the emitter voltage “E” divided by the external resistance Rextvalue “R”, or

I=E/R  [Ohm's law]

where “I” is the PTAT current I(T). The rate of change with temperature“T” of the current I(T) is a slope determined by taking the derivativeof this Ohm's law equation: $\begin{matrix}{\frac{I}{T} = {{\frac{1}{R}\frac{E}{T}} - {\frac{E}{R^{2}}\frac{R}{T}}}} & \lbrack {{EQ}.\quad C} \rbrack\end{matrix}$

If the external resistance Rext is a fixed resistor, then the secondterm in EQ. C nulls out (dR/dT equals zero) and the rate of currentchange with temperature dI/dT is directly proportional to the rate ofchange of the emitter voltage dE/dT, with the proportionality constantbeing the fixed value 1/R. Therefore the slope dI/dT can be increased bydecreasing the value of R, but from Ohm's law we see that such a changewill also increase the level of the current I (the PTAT current I(T)).It should be noted that the sign of the slope dI/dT is irrelevant tothis discussion, since whatever sign results can be accommodated bycalibration constants used in calculating temperature from the output ofthe temperature sensing circuit 306 or of a transponder 200, 400 whichutilizes the temperature sensing circuit 306. Thus, for example, a“steeper” or “increased” slope herein refers to any slope which has anabsolute value greater than that of another slope.

In its simplest form, the present invention utilizes a thermistor 716for the external resistance Rext. Since thermistors have a resistancewhich predictably varies with temperature, the slope dI/dT is influencedby the second term in EQ. C. Furthermore, if the nominal value ofresistance for the thermistor 716 is similar to the value of thepreviously-used fixed resistor 216, 416, then the PTAT current I(T) willremain close to the same current level as before (due to Ohm's law). Fortesting of the best mode embodiment of the invention, a negativetemperature coefficient (NTC type) thermistor was selected, such asThermometric Corporation of Edison, New Jersey's part # BR42KB504M whichhas a nominal “zero-power” resistance of 500 kilohms at 25° C.(Subsequent to the testing it was decided that a better model ofthermistor in terms of packaging and tolerance may be a part #NHQMM504B435R5.) A negative temperature coefficient means that dR/dT isnegative, so that when the second term in equation EQ. C is subtractedfrom the first term, the effect is to provide a positive addition to theslope dI/dT. Since dE/dT is negative making the first term negative, theslope dI/dT can be changed from negative to less negative, to zero, orto positive depending on the relative magnitudes of the first and secondterms in equation EQ. C. Finally, it should be noted that NTCthermistors have resistance which varies with temperature in anon-linear way (e.g., approximately proportional to exp(1/T)), howeverover a sufficiently small temperature range a linear approximationsuffices.

FIG. 6 is a graph 690 which illustrates the thermistor test resultscompared with linear approximations of the thermistor results and of thefixed resistor results, all implemented in a transponder 400 (comparableto 200). Referring to FIG. 6, the vertical axis 694 is the temperaturecount N_(T) (proportional to the PTAT current I(T)) which is plottedversus the temperature T (measured in degrees Fahrenheit) on thehorizontal axis 692. The temperature range (90° F.−110° F.) for the testwas selected because it encompasses temperatures suitable for bodytemperature measurements. The thermistor test (utilizing the circuit700) produced the curve 698 which is then approximated by a straightline curve 699 which is curve-fit to the middle portion of the curve698. The horizontal flat spots in the curve 699 are due to noise in thetest setup, but even ignoring them, the inherent curvature due to thethermistor's nonlinearity can be observed. However, it can also be seenthat for the range of 95° F.−105° F. (35° C.−41° C.) the linearapproximation of the straight line curve 699 is quite good. Such a rangeis adequate for body temperature measurements. For comparison purposes,a straight line curve 696 is drawn in to represent the linearizedresults of past tests with a fixed resistor Rext having a resistance of453 kilohms. The fixed resistor curve 696 has a negative slope of −3.89counts/° F. or a resolution of 0.26° F. (per N_(T) count). Thelinearized thermistor curve 699 has a positive slope of 57.8 counts/° F.or a resolution of 0.02° F. The inventive use of the thermistor 716 inplace of a fixed resistor 216, 416 for the external resistance Rext hasproduced a 15-fold improvement in resolution for temperaturemeasurements made by the temperature sensing circuit 306. The testedthermistor 716 has a resistance of 312 kilohms at 95° F. (35° C.) and248 kilohms at 104° F. (40° C.) which are above the minimum allowedresistance of 200 kilohms for the transponder 400. The same slope (andresolution) could be obtained with resistance values around 500 kilohmsby simply substituting a thermistor of the same design with a zero-powerresistance of 800 kilohms −900 kilohms at 25° C., such as theThermometrics part # BR42KB824M. By way of comparison, if a fixedresistor were to be used to obtain the same (although negative) slope asfor the thermistor, a calculation utilizing the first term in equationEQ. C tells us that a 30 kilohm fixed resistor 216, 416 would have to beused. This would yield unacceptably high PTAT current I(T) levels in thetransponder 400 which is limited to currents below those produced by aminimum external resistance Rext value of 200 kilohms.

As mentioned hereinabove, a possible outcome of utilizing an NTCthermistor 716 is to counterbalance the transistor Q1 temperatureresponse, thereby producing an approximately null temperature response,i.e., a slope of approximately zero counts per degree. Since thetransistor Q1 response is substantially linear over a wide temperaturerange, and the thermistor response is only approximately linear over arelatively narrow temperature range, this null response would only holdapproximately true for the thermistor's narrow “linear” temperaturerange. Nevertheless, for transponders such as the transponders 200 and400 described herein which measure both temperature and pressure, butproduce a pressure-indicative signal which is confounded with asimultaneous temperature response, nulling out the temperature responsewould allow the transponder to be used to measure pressure only whileignoring the temperature, at least within a narrow temperature range.Alternatively, such a nulled response temperature measuring device couldbe used in a situation wherein a step response, or non-zero response isdesired as an attribute indicator that the measured temperature iseither “in” (zero response) or “out” (non-zero response) of a desiredtemperature range. No doubt a reader skilled in the relevant arts candetermine other uses, all of which are intended to be within the scopeof the present invention.

The scope of the invention comprising the temperature sensing circuit306 with external resistance Rext 216, 416, 716 is intended to includeconstructions wherein the “external” resistance Rext is provided byresistances, including temperature-responsive resistances such as theabovementioned thermistor 716, which are at least partially implementedintegrally in the IC chip 202, 402. For example, the resistance Rext canbe provided by polysilicon resistor construction. In this sense, theresistance Rext remains external to the temperature sensing circuit 306,but not to the IC chip 202, 402.

Another embodiment of the transponder 400 will now be described. Thisembodiment provides a way to overcome a potential problem with“rollover” of the transponder output count (temperature reading N_(T)).The rollover problem is illustrated by FIG. 8 which is a graph 890 ofcount (e.g., temperature reading count N_(T) or pressure reading countN_(P)) on the vertical axis 894, versus temperature in degrees Celsiuson the horizontal axis 892. A temperature response curve 898 hasreadings with count values which increase monotonically with temperatureuntil just after reaching a highest point 881, and then “rolls over” toa lowest point 883 before resuming a monotonic increase withtemperature. The rollover occurs because the data capture circuit 254,454 has a counter which is limited to, in this example, 12 bits or acount of 4095. As is known for 12 bit counters, adding one more count to4095 results in clearing all 12 bits to produce a count of zero. Asdiscussed hereinabove, by varying the resistance value of the externalresistor Rext 216, 416, 716, and also by varying the characteristics ofthe current scaling circuit 310, 510 (e.g., by setting the trimming bits436 b) a transponder user can attempt to avoid rollovers within thetemperature and/or pressure range of interest to the user. However, insituations such as described immediately above, wherein the transponderhas been modified in order to produce a steeper slope in the temperatureresponse, it becomes even more difficult to prevent rollover since thesteeper slope makes it more likely that the highest point 881 will bereached within the temperature range of interest (e.g., 34-48° C.).Given the limited range of available values for thermistor nominalresistance, the restrictions on Rext resistance values due totransponder current-handling limitations, and the limited range for trimadjustments of the current scaling (especially for the temperaturemeasuring mode of operation), it may not be possible to avoid rolloverwithin certain desired temperature measurement ranges. The inventivesolution to this problem is to utilize the pressure measurement mode ofoperation for measuring temperature by replacing the external measuringcapacitor C_(P), 418 with a fixed capacitor having a capacitance valuewhich is substantially independent of both temperature and pressure.Since the measuring capacitor C_(P), 418 no longer has apressure-sensing capacitance which varies with pressure, the ability ofthe transponder 400 to measure pressure is sacrificed, but now a newvariable has been introduced for easily adjusting the output reading.Since the measuring capacitor C_(P), 418 is external to the transponder400, it is easily changed. With a fixed value measuring capacitor C_(P),418, the “pressure” output reading N_(P) which is determined during thepressure measurement mode of transponder operation, will actually varyonly with temperature, due to the temperature variance of themeasurement current I(T)B. Thus the pressure count N_(P) measurementscan be used to form a temperature response curve 897 (see FIG. 8) whichdoes not rollover, even though the temperature count N_(T) measurementsform a temperature response curve 898 which does rollover. Thus, bysuitable selection of a fixed value for the capacitance of the externalmeasuring capacitor C_(P), 418, and of a value for the scaling factortrimming bits, the pressure reading N_(P) plotted versus temperature hasa monotonic slope for all values of temperature within a desiredtemperature measurement range. Greater adjustment flexibility comesabout since the capacitance value of the measuring capacitor C_(P), 418can be adjusted without concern for current limitations, and a widerange of capacitance values are available for suitable externalcapacitors. Also, in the transponder 400 design, there is a wider rangeof current scaling available during the pressure measurement mode ofoperation.

Referring to FIG. 5, it can be seen that an exemplary way to produce thetwo response curves 897 and 898 (as shown in FIG. 8) is to select afixed capacitance value for the capacitor C_(P), 418 which is greaterthan the fixed capacitance value for the capacitor C_(FX). Forsimplicity of this example, assume that the current scaling factor B isthe same for both pressure and temperature measurement mode operation(e.g., the trimming bits TRIMBIT_5 through TRIMBIT_9 are all cleared sothat the current scaling factor B is 1.0 regardless of mode). Nowcompared to the (smaller valued) C_(FX) capacitor, a given measurementcurrent I(T)B will take longer to charge/uncharge the C_(P) capacitor tothe triggering voltage levels (e.g., Vbg and Vbg/2) for the set-resetportion 514 of the relaxation oscillator 512, thereby producing a lowerfrequency output signal Fosc′ and thus a count N_(P) which is lower thanthe count N_(T). The temperature variation remains in both responsecurves 897, 898 because of the temperature dependence of the measurementcurrent I(T)B. If the wider range of current scaling factors B is to beutilized in the “pressure” measurement mode, the larger resultingcurrents I(T)B will have to be balanced off by suitably largercapacitance values for the measuring capacitor C_(P), 418.

It should be noted that the beneficial effects of utilizing a fixedvalue measuring capacitor C_(P), 418 for temperature measurement may beobtained whether or not a thermistor 716 is utilized for the resistanceRext; however, the use of the thermistor 716 may make the employment ofthe fixed value measuring capacitor C_(P), 418 necessary to avoidrollover.

Finally, it should be noted that the same inventive concept of utilizinga fixed value external measuring capacitor C_(P) for temperaturemeasurement can be employed with the previous model of transponder 200.Referring to FIG. 3, it can be seen that if the measuring capacitorC_(P), 218 is replaced with a fixed capacitor having a capacitance valuewhich is substantially independent of both temperature and pressure,then during the pressure measurement mode when the capacitor C_(P), 218is connected by the switch 350, the measuring capacitance(C_(FX2)+C_(P)) is automatically greater than during the temperaturemeasurement mode, thereby producing lower counts N_(P) compared to thecounts N_(T).

Although the invention has been illustrated and described in detail inthe drawings and foregoing description, the same is to be considered asillustrative and not restrictive in character, it being understood thatonly preferred embodiments have been shown and described, and that allchanges and modifications that come within the scope of the inventionare desired to be protected. Undoubtedly, many other “variations” on the“themes” set forth hereinabove will occur to one having ordinary skillin the art to which the present invention most nearly pertains, and suchvariations are intended to be within the scope of the invention, asdisclosed herein.

What is claimed is:
 1. A temperature measurement device, characterizedby: a resistance (Rext); and a temperature sensing circuit comprising atemperature sensing transistor which exhibits a predictable change inits base-emitter voltage due to temperature, and transistors connectedfor mirroring a current (I(T)) through the temperature sensingtransistor and through the resistance.
 2. A temperature measurementdevice according to claim 1, further characterized in that: theresistance is a fixed resistor.
 3. A temperature measurement deviceaccording to claim 2, further characterized in that: the fixed resistorhas a resistance value which is substantially independent oftemperature.
 4. A temperature measurement device according to claim 3,further characterized in that: the resistance value is a fixed valuebetween about 20.5 kilohms and about 455 kilohms.
 5. A temperaturemeasurement device according to claim 3, further characterized in that:the resistance value is a fixed value greater than about 200 kilohms. 6.A temperature measurement device according to claim 1, furthercharacterized in that: the resistance has a resistance value whichpredictably varies with temperature.
 7. A temperature measurement deviceaccording to claim 6, further characterized in that: the resistance is athermistor.
 8. A temperature measurement device according to claim 6,further characterized in that: within a desired temperature measurementrange, the resistance has a resistance value great enough to preventunacceptable levels of the current.
 9. A method of adjusting temperatureresponse for a temperature sensing circuit connected to avoltage-to-current converting resistance (Rext) wherein the temperaturesensing circuit includes a temperature sensing transistor which exhibitsa predictable change in its base-emitter voltage due to temperature, andtransistors connected for mirroring a current through the temperaturesensing transistor and through the resistance, characterized by the stepof: selecting the resistance value in order to produce a desired slopefor the temperature response.
 10. A method according to claim 9,characterized by: utilizing a fixed resistor for the resistance.
 11. Amethod according to claim 9, characterized by: utilizing a resistancethat predictably varies with temperature.
 12. A method according toclaim 11, characterized by: selecting a temperature coefficient for theresistance that has a value great enough compared to the temperaturecoefficient of the temperature sensing transistor in order to increasethe slope of the temperature response compared to the slope that wouldresult from utilizing a minimum resistance valued fixed resistor for theresistance; and selecting a nominal value for the resistance such thatthe resistance values are great enough to prevent unacceptable levels ofthe current within a desired temperature measurement range.
 13. A methodaccording to claim 11, characterized by: selecting a temperaturecoefficient for the resistance that counterbalances the temperaturecoefficient of the temperature sensing transistor in order to produce anapproximately zero slope of the temperature response within a portion ofa desired temperature measurement range; and selecting a nominal valuefor the resistance such that the temperature-varying resistance value isgreat enough to prevent unacceptable levels of the current within thedesired temperature measurement range.
 14. An RF transponder,characterized by: a resistance (Rext); a temperature sensing circuitcomprising a temperature sensing transistor that exhibits a predictablechange in its base-emitter voltage due to temperature, and transistorsconnected for mirroring a temperature-indicative current through thetemperature sensing transistor and through the resistance; circuitry forconverting the mirrored current to a temperature reading that isproportional to the mirrored current; and a value for the resistancethat predictably varies with temperature.
 15. An RF transponderaccording to claim 14, further characterized in that: the resistance isa thermistor.
 16. An RF transponder according to claim 14, furthercharacterized in that: within a desired temperature measurement range,the resistance has resistance values great enough to preventunacceptable levels of the mirrored current.
 17. An RF transponderaccording to claim 16, further characterized in that: the resistance hasa temperature coefficient great enough, relative to the temperaturecoefficient of the temperature sensing transistor, to increase the slopeof the temperature reading versus temperature compared to the slope thatwould result from utilizing a minimum resistance valued fixed resistorfor the resistance.
 18. An RF transponder according to claim 16, furthercharacterized in that: the resistance has a temperature coefficientwhich counterbalances the temperature coefficient of the temperaturesensing transistor in order to produce an approximately zero slope ofthe temperature reading versus temperature within a portion of a desiredtemperature measurement range.
 19. An RF transponder, characterized by:a resistance (Rext); a temperature sensing circuit comprising atemperature sensing transistor which exhibits a predictable change inits base-emitter voltage due to temperature, and transistors connectedfor mirroring a temperature-indicative current through the temperaturesensing transistor and through the resistance; an external measuringcapacitor having a capacitance value which is fixed and substantiallyindependent of temperature and pressure; a relaxation oscillator circuitwhich utilizes the external measuring capacitor to convert thetemperature-indicative current to an output signal; and a data capturecircuit for converting the output signal to a reading which isproportional to the temperature-indicative current.
 20. An RFtransponder according to claim 19, further characterized in that: thevalue for the capacitance of the external measuring capacitor isselected so that the reading plotted versus temperature has a monotonicslope for all values of temperature within a desired temperaturemeasurement range.
 21. A method of adjusting temperature response for atemperature sensing circuit connected to a voltage-to-current convertingresistance (Rext) wherein the temperature sensing circuit includes atemperature sensing transistor which exhibits a predictable change inits base-emitter voltage due to temperature, and transistors connectedfor mirroring a current through the temperature sensing transistor andthrough the resistance, characterized by the steps of: selecting theresistance value in order to produce a desired slope for the temperatureresponse; utilizing a resistance that predictably varies withtemperature; selecting a temperature coefficient for the resistance thathas a value great enough compared to the temperature coefficient of thetemperature sensing transistor in order to increase the slope of thetemperature response compared to the slope that would result fromutilizing a minimum resistance valued fixed resistor for the resistance;and selecting a nominal value for the resistance such that theresistance values are great enough to prevent unacceptable levels of thecurrent within a desired temperature measurement range.
 22. A method ofadjusting temperature response for a temperature sensing circuitconnected to a voltage-to-current converting resistance (Rext) whereinthe temperature sensing circuit includes a temperature sensingtransistor which exhibits a predictable change in its base-emittervoltage due to temperature, and transistors connected for mirroring acurrent through the temperature sensing transistor and through theresistance, characterized by the steps of: selecting the resistancevalue in order to produce a desired slope for the temperature response;utilizing a resistance that predictably varies with temperature;selecting a temperature coefficient for the resistance thatcounterbalances the temperature coefficient of the temperature sensingtransistor in order to produce an approximately zero slope of thetemperature response within a portion of a desired temperaturemeasurement range; and selecting a nominal value for the resistance suchthat the temperature-varying resistance value is great enough to preventunacceptable levels of the current within the desired temperaturemeasurement range.
 23. An RF transponder, characterized by: a resistance(Rext); a temperature sensing circuit comprising a temperature sensingtransistor that exhibits a predictable change in its base-emittervoltage due to temperature, and transistors connected for mirroring atemperature-indicative current through the temperature sensingtransistor and through the resistance; circuitry for converting themirrored current to a temperature reading that is proportional to themirrored current; a value for the resistance that predictably varieswith temperature; such that within a desired temperature measurementrange, the resistance has resistance values great enough to preventunacceptable levels of the mirrored current; and the resistance has atemperature coefficient great enough, relative to the temperaturecoefficient of the temperature sensing transistor, to increase the slopeof the temperature reading versus temperature compared to the slope thatwould result from utilizing a minimum resistance valued fixed resistorfor the resistance.
 24. An RF transponder, characterized by: aresistance (Rext); a temperature sensing circuit comprising atemperature sensing transistor that exhibits a predictable change in itsbase-emitter voltage due to temperature, and transistors connected formirroring a temperature-indicative current through the temperaturesensing transistor and through the resistance; circuitry for convertingthe mirrored current to a temperature reading that is proportional tothe mirrored current; a value for the resistance that predictably varieswith temperature; such that: within a desired temperature measurementrange, the resistance has resistance values great enough to preventunacceptable levels of the mirrored current; and the resistance has atemperature coefficient which counterbalances the temperaturecoefficient of the temperature sensing transistor in order to produce anapproximately zero slope of the temperature reading versus temperaturewithin a portion of a desired temperature measurement range.